Sensorless Position PIM

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 9, SEPTEMBER 2011 3815

Rotor Design for Sensorless Position Estimation inPermanent-Magnet Machines

Rafal Wrobel, Alan S. Budden, Dan Salt, Derrick Holliday, Phil H. Mellor,Andrei Dinu, Parmider Sangha, and Mark Holme

Abstract—A high-frequency injection sensorless rotor positionestimation algorithm is incorporated directly into the finite-element design process to realize a permanent-magnet (PM) ma-chine that is suited to zero- and low-speed sensorless control.The machine design is tightly constrained by an existing statorassembly, only enabling the redesign of the replacement PM rotor,and by the requirement that the manufacture of the resultingmachine must be simple while meeting aerospace standards. Ex-perimental results confirm that the resulting machine matchesthe original performance specification and that it operates underposition sensorless control.

Index Terms—Permanent-magnet (PM) machine, sensorlesscontrol.

I. INTRODUCTION

B RUSHLESS permanent-magnet (PM) machines offer apower dense actuation technology that is increasingly

being deployed in “more electric” applications in the aerospaceand automotive sectors. Correct commutation of the machinephases requires knowledge of the angular position of the PMrotor which, for brushless dc operation, is usually achieved us-ing Hall-effect transducers. For high-performance brushless acoperation, a resolver or an encoder is typically required whichmay add considerable cost to the drive system while reducing itsruggedness. One method of overcoming these disadvantages isto use sensorless control, where the rotor position is estimatedby applying signal processing techniques to measurements ofthe machine’s voltages and currents, obtained using low-cost,rugged, and remotely positioned transducers that are normallyalready included in the drive system for current control strate-gies or fault detection [1].

Sensorless position estimation in PM machines is reliantupon the existence of a measurable electrical parameter thatis related to rotor position. When the machine is operating

Manuscript received April 16, 2010; revised August 26, 2010 andOctober 25, 2010; accepted October 28, 2010. Date of publicationNovember 18, 2010; date of current version August 12, 2011.

R. Wrobel, D. Salt, D. Holliday, and P. H. Mellor are with the Universityof Bristol, BS8 1TH Bristol, U.K. (e-mail: [email protected];[email protected]; [email protected]; [email protected]).

A. S. Budden is with the Controls Division, Rotork Controls Ltd., Bath BA13JQ, U.K. (e-mail: [email protected]).

A. Dinu, P. Sangha, and M. Holme are with Engine Control Systems,Goodrich Corporation, Birmingham B28 8LN, U.K. (e-mail: [email protected]; [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2010.2093484

at speed, the electromotive force (EMF) generated across themachine supply terminals provides a suitable rotor-position-dependent quantity [2]–[6]. When the rotor is stationary oroperating at low speed, however, no rotor-position-dependentparameter is readily available, and more complex methodsthat typically depend upon detection of the machine’s rotor-dependent inductance profile are used [7]–[19].

Typically, a PM machine may exhibit some degree ofmagnetic saliency which will cause a rotor-position-dependentvariation in the inductance of the stator windings.

This can be detected by the injection of a low-magnitudehigh-frequency voltage set into the stator and the measurementof the resulting currents [10], [13]. The degree of saliency, andtherefore the resulting inductance variation, is highly dependentupon machine design, and comparisons between different PMmachines have shown that, under specific operating conditions,certain rotor topologies and saliency profiles are more suitedto sensorless control than others [20]. The rotor topology influ-ences the selection of an appropriate injection-based sensorlesscontrol strategy, particularly for use under high-load conditions[21], and rotor geometry has significant influence on the accu-racy of the estimation method [22], [23].

To capitalize on the advantages of sensorless control, there-fore, consideration may be given to the saliency profile at thedesign stage of the PM machine. Design methodologies for“flux bridges” that introduce measurable saliencies in struc-turally symmetrical machines, at the expense of a relativelycomplex and difficult-to-manufacture rotor structure, have beenproposed [24]. Finite-element (FE) and experimental studiesof the effects of magnetic saturation and cross-coupling be-tween the d- and q-axis fluxes on the accuracy of injection-based sensorless control in machines with interior magnet rotortopologies have resulted in design recommendations relating torotor geometry, the magnitude of the PM flux, and the relativemagnitudes of the d- and q-axis fluxes [25]–[27].

In summary, therefore, multislot distributed stator windingsprovide a clear saliency profile when compared to concentratedwound machines where slotting effects are significant. Conse-quently, there is a compromise between the benefits of torquedense rugged machine construction and the “ideal” design whena measurable saliency profile to enable rotor position detectionand simplicity of manufacture are both major considerations.

The design of a PM rotor for a concentrated wound torquedense brushless dc machine that exhibits a saliency profilesuited to voltage-injection-based sensorless control is pre-sented. The design methodology directly combines a sensorlessposition estimation algorithm with FE analysis [28]–[30]. The

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3816 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 58, NO. 9, SEPTEMBER 2011

Fig. 1. Baseline concentrated wound surface magnet topology.

design is constrained by the requirements for the rotor to fitwithin an existing stator, to meet stringent aerospace designstandards, and for sensorless position estimation to operate atzero and low speeds where the torque demand does not exceed0.5 p.u. Particular consideration is given to choice of the dimen-sions of the PM segments to result in a usable saliency profilewhile maintaining low torque ripple. The method is illustratedfor the case where the machine is unloaded and does not in-corporate saliency variation resulting from saturation, althoughthe method may readily be extended to include load currentas an additional design parameter. The effect of saturation onthe machine incorporating the new rotor is however illustratedusing FE analysis. Mechanical ruggedness and simplicity ofmanufacture are also key design considerations.

II. DESIGN CONSIDERATIONS

A prototype PM machine had previously been designed andoptimized to meet stringent power density and performancerequirements. Specifically, the machine is designed for deploy-ment in an aerospace actuator application requiring high peak-to-average torque ratio and low operating duty. This machinecomprises a concentrated wound stator with 1.5 slots per poleand a uniformly magnetized rotor with NdFeB magnet seg-ments bonded to the surface of a magnetically permeable steelhub, as shown in Fig. 1. The magnets are further retained usinga metallic sleeve that encapsulates the rotor.

The saliency exhibited by the “baseline” surface magnetmachine is insufficient for the new requirement that it shouldoperate under position sensorless control. The interior PM(IPM) rotor topology exhibits more favorable saliency charac-teristics [31], [32] and was therefore chosen as the basis for thedevelopment of a new rotor.

Importantly, while having a saliency profile suited toinjection-based sensorless control, the new rotor must also bea direct physical replacement for the existing surface-mountedPM rotor, must match the torque capability of the existingbaseline machine, and must be compatible with establishedaerospace manufacturing practice.

A. Manufacturing Considerations

The rotor generally consists of a solid magnetically per-meable back-iron structure profiled to accept the PMs and

Fig. 2. Proposed IPM arrangement.

fixed to the rotor shaft. To simplify manufacture, the proposedrotor would use soft magnetic composite (SMC) pole pieceswith rectangular PMs arranged in between a “spoke” pattern.This offers a quick and simple assembly process and reducedcomplexity when compared to designs that use laminations.

The proposed rotor design is shown in Fig. 2, where thecomposite pole pieces and PMs are bonded to a nonmagneticsteel hub and the structure is secured using a thin sleeve.

B. Sensorless Control Algorithm

The high-frequency injection-based sensorless algorithm isa key influence on the design of the new rotor. This sectiontherefore describes the sensorless technique that is consideredin combination with the FE techniques of Section III during therotor design process.

The standard dq model for a PM machine is shown in

vdq =(

Rs + pLd ωrLq

ωrLd Rs + pLq

)idq +

(0

ψmωr

). (1)

If the angular frequency ωi of the injected voltage is suffi-ciently high, the stator resistance may be considered negligiblein comparison with the stator reactance. Equation (2) thereforeexpresses the resulting approximate high-frequency machinemodel in the stationary αβ reference frame. The inductanceterms Lαβ are obtained by transforming the dq-axis induc-tances into the two-phase stationary reference frame(

vα

vβ

)≈

(jωiLαiαjωiLβiβ

). (2)

If the balanced high-frequency voltage set shown in (3) isinjected into the model of (2), the resulting high-frequencycurrent can be calculated as shown in (4), where I1 and I2 aredependent upon the machine inductance and the magnitude andfrequency of the injected voltage set [33]

vαβ =Vi

(− sin(ωit)cos(ωit)

)(3)

iαβ = I1

(cos(ωit)sin(ωit)

)− I2

(cos(2θr − ωit)sin(2θr − ωit)

). (4)

The rotor position information θr contained in the secondterm of (4) is obtained by using the Park transform to rotate

WROBEL et al.: ROTOR DESIGN FOR SENSORLESS POSITION ESTIMATION IN PERMANENT-MAGNET MACHINES 3817

Fig. 3. Sensorless rotor position estimation scheme.

the αβ currents to a reference frame that is synchronous withthe injected voltage set. This also has the effect of introducingcurrent terms rotating at twice the injected frequency. High-passfiltering then results in the currents given in(

i′xi′y

)= −I2

(cos(2θr − 2ωit)sin(2θr − 2ωit)

). (5)

Finally, applying the Park transform to rotate the terms of (5)through an angle of θ = −2ωit and then low-pass filteringresult in a signal in which angle is directly related to rotorposition, as shown in(

i′′xi′′y

)= −I2

(cos(2θr)sin(2θr)

). (6)

A simple arctangent calculation or a tracking observer [11],[33] may be used to complete the sensorless scheme which issummarized in Fig. 3.

For machines with sinusoidal saliency profiles, plotting i′′xversus i′′y results in a circular locus or “saliency fingerprint.”In practical machines, however, the saliency fingerprint rarelyfollows this fixed circular profile, as a result of slotting and sat-uration for example, and this introduces error to the estimatedrotor position. This error can be compensated using a trackingobserver or lookup-table techniques [11], [33].

III. FE DESIGN

The manufacturing considerations described in Section II-Adictate that torque and magnetic saliency are key design para-meters. In addition, the predefined stator structure and airgapand the proposed rotor structure and materials limit the designvariables to the depth and width of the rectangular magnetsegments.

A. FE Modeling

The FE method is used to calculate the torque and magneticflux linkage of the phase windings at discrete rotor positionsover an electrical cycle. Since the proposed IPM rotor machineexhibits radial flux, analysis can be carried out using 2-D FEmodeling techniques with the calculation area being limited tothe machine’s cross section. For simplicity, end effects havebeen neglected: The validity of this approach is justified bythe accuracy of the experimental results presented in Section V.Due to the periodicity shown in Figs. 1 and 2, the analysis only

requires consideration of two pole pitches. To simulate rotationof the rotor, the moving-band technique is used, and the solverused in the FEM analysis considers the magnetostatic field andaccounts for material nonlinearity.

The magnet depth is expressed in terms of the ratio of theinner dimension Rm to the outer radius Rr, and the width isexpressed as the ratio of an angle γm subtended by the magnetto the pole pitch γp, as shown in Fig. 2. These quantities arerepresented in the set of dimensionless parameters x, as shownin (7), which is used to specify the motor structure used in theFEM analysis

x =[γm

γp,Rm

Rr

]. (7)

Torque is calculated using the coenergy method shown in (8),where W ′ is the magnetic coenergy, Θ is the angle defining therelative position between the rotor and the stator, and ΔΘ is theangular rotation between field solutions

T =∂W ′

∂Θ

∣∣∣∣I=const

=W ′(Θ) − W ′(Θ − ΔΘ)

ΔΘ

∣∣∣∣I=const

. (8)

The calculated torque is averaged over successive 60◦eleccommutations of a (0,+1,−1) Irated winding current patternto emulate brushless dc operation. Saliency torque is accountedfor by adjusting the commutation angle to yield a maximumaverage torque. The average torque Ti developed by the ithmotor version is normalized to the average constant torque Tproduced by the baseline prototype machine.

The magnetic flux linkage for a single coil of the statorwinding is found using (9), where A is the magnetic vectorpotential (i.e., A = Az), nt is the number of turns in a singlecoil, l is the active length of the machine, and S is the cross-sectional area of a coil for positive or negative current flowonly (S = S+ = S−). To derive the flux linkage for a completestator phase winding, the number of coils per phase must beincorporated into

ψ =ntl

S

⎛⎜⎝

∫∫S+

AdS −∫∫S−

AdS

⎞⎟⎠ . (9)

Based on the magnetic flux linkage of (9), the saliency ratioχi for the ith motor version, defined by (10), where Ld and Lq

are the d- and q-axis inductances, respectively, is calculated.Note that (10) is the inverse of the more usual definition


Fig. 4. Distribution of the relative average torque versus γm/γp andRm/Rr .

Fig. 5. Distribution of the saliency ratio versus γm/γp and Rm/Rr .

of the saliency ratio and reflects the inverse saliency that ischaracteristic of the IPM machine

χi =(Lq)i

(Ld)i. (10)

Both the torque and saliency ratio calculations are re-peated for different values of x, with target values ofTi/T = 1 to ensure the correct torque performance and χi =1.1, 1.2, 1.3, . . . , 2.5 to ensure a degree of rotor saliency that issuitable for sensorless control.

The FE analysis incorporates an objective function, definedin (11), which seeks to minimize the error between the calcu-lated and target values of torque and saliency

f(x) = minx∈�n

(∣∣∣∣Ti

T− 1

∣∣∣∣ +∣∣∣∣χi

χ− 1

∣∣∣∣)

. (11)

The normalized average torque and saliency ratio are bothmapped in a 2-D solution space for discrete steps of γm/γp andRm/Rr, as shown in Figs. 4 and 5.

Fig. 4 shows that, in general, the torque developed by themotor increases as γm/γp (magnet width) increases and asRm/Rr (magnet depth) decreases. The saliency surface, shown

TABLE IROTOR DESIGN DATA

Fig. 6. FE model for rotor version 3.

in Fig. 5, is more complex and shows no simple trend in relationto changes to γm/γp and Rm/Rr.

It is important to note that, to simplify the presentation ofthe design technique, the results shown in Figs. 4 and 5 arefor an unloaded machine. However, the method may be easilyextended to include specific load conditions, defined by thestator current vector, and a perturbation to represent the signalinjection used in the sensorless technique. In doing this, cross-coupling and saturation effects are captured in the analysis.

Table I presents the target torque and saliency ratio valuesand the corresponding values resulting from the FE designprocess which are also highlighted on the surface plots ofFigs. 4 and 5. It is important to note that the saliency ratiosshown in Table I result from calculation at a fixed rotor positionand that, beyond the limits of γm/γp = 0.73 and γm/γp =0.93, none of the resulting rotor versions meets the designspecification.

Fig. 6 shows the FE model, together with the mesh dis-cretization, for rotor version 3. Due to symmetry, the 2-D modelincludes only one-third of the motor’s cross section.

The 14 rotor versions shown in Table I, resulting from themanipulation of the PM segment dimensions, are all capable ofmeeting the design specification and were therefore subject tomore detailed investigation.


Fig. 7. Saliency analysis incorporated into the FE design procedure.

B. Saliency Analysis

Ideally, in order to assess the suitability of a rotor design forzero- and low-speed sensorless position estimation, an accuratemachine model would be combined with the voltage injectionsystem and sensorless algorithm. The computational complex-ity of modeling the injection scheme prohibits the use of thisdirect approach. It has been shown, however, that the behaviorof the sensorless scheme can be accurately reproduced usingFE techniques by incorporating a suitable three-phase currentpattern and calculating the resulting flux linkage in the machinewinding [34].

For each of the 14 candidate rotor designs shown in Table I,FE simulations were carried out, where the machine windingswere energized using a balanced three-phase set of sinusoidalcurrents superimposed upon the dc excitation current. Thethree-phase current set is defined by (12), where θi is the angleof the injection signal for a given sample and is equivalent toωit in the practical system described in Section II-B

⎛⎝ Ia

Ib

Ic

⎞⎠ = Ipk

⎛⎝ sin(θi)

sin(θi − 2π/3)sin(θi + 2π/3)

⎞⎠ . (12)

Using the FE postprocessor, the corresponding flux linkagesψa, ψb, and ψc of the stator phase coils are found from thevector potential solution for each rotor angle, as described by(9). The computational overhead is minimized by consideringthe symmetry of the results which enables calculation to becarried out over only one-sixth of an electrical cycle. Theflux linkage that is attributable to the three-phase current setonly, without any contribution from the fundamental excitationcurrent or PMs, is found using (13), where ψinj is the phase fluxlinkage including the injected three-phase set and ψm is the fluxlinkage without signal injection

ψ(θi)|Θ=const = ψinj(θi)|Θ=const − ψm|Θ=const. (13)

The flux linkages are processed in the same way as the prac-tical sensorless rotor position estimation scheme, and saliencyfingerprints for a given rotor topology can be produced. Theprocess is summarized in Fig. 7, which is equivalent to Fig. 3,where ψαβ is obtained by applying the Clarke transform to theflux linkages ψabc obtained from the FE process.

Fig. 8 shows a saliency fingerprint for rotor version 1, whosecharacteristics are defined in Table I, obtained using FE tech-niques. The ideal circular profile shows sinusoidally varying

Fig. 8. Saliency fingerprint for rotor version 1.

saliency, while the inner “distorted triangular” profile describesthe saliency of rotor version 1. The lines extending fromthe center of the plot, through the saliency fingerprint to theideal circular saliency characteristic, show the relationship be-tween the two plots at evenly spaced rotor positions and demon-strate the tendency for the estimated rotor positions to clusteraround the apexes of the saliency fingerprint.

Fig. 9 shows the corresponding estimated rotor position forrotor version 1, obtained using FE techniques, and highlightsthe significant distortion resulting from the nonideal saliencyprofile.

The analysis shown in Figs. 8 and 9 was repeated for all14 rotor versions defined in Table I. Figs. 10 and 11 show theresulting saliency fingerprint and estimated rotor position forrotor version 14 and highlight the significant distortion of thesaliency fingerprint and the consequent reduction in positionaccuracy as the PM dimensions are varied.

The maximum position error over one complete electricalcycle, commonly used as a measure of the effectiveness ofsensorless position estimation strategies, was calculated foreach rotor version to enable identification of the best candidatedesign and comparison with other published sensorless tech-niques. The calculated maximum errors, expressed in electricalradians, for all 14 rotor versions are summarized in Fig. 12. Thefigure shows that rotor versions 1–3 provide the most accuratesensorless position estimate and are therefore the most likelycandidates for further development.


Fig. 9. Estimated rotor position for rotor version 1.

Fig. 10. Saliency fingerprint for rotor version 14.

Fig. 11. Estimated rotor position for rotor version 14.

Fig. 12. Maximum error for rotor versions 1–14.

Fig. 13. Variation in relative peak cogging torque.

C. Cogging Torque

As well as the baseline torque and saliency requirements,the application for the new rotor design demands low coggingtorque. FE analysis was used to investigate the cogging exhib-ited by each of the 14 candidate rotor designs.

The results, which are shown in Fig. 13, show that rotorversions 3–7 all exhibit relatively low cogging torque.

The design process outlined in Section III-A–C shows that, incomparison with the candidate rotor designs shown in Table I,rotor version 3 exhibits the best combination of torque capa-bility, saliency, position estimation accuracy, and low coggingtorque. Rotor version 3 was therefore selected for manufacture.

IV. MANUFACTURE OF ROTOR VERSION 3

Fig. 14 shows the construction of rotor version 3. Fig. 14(a)shows the simple structure consisting of the shaft and hubassembly, the PM segments prior to magnetization, and theSMC pole pieces. Both the SMC pole pieces and PM segmentswere machined to shape using the same technique, whichsignificantly simplifies the manufacture and assembly of the


Fig. 14. IPM rotor version 3. (a) Rotor components. (b) Assembled rotor priorto the addition of the containment sleeve.

rotor. Fig. 14(b) shows the assembled rotor, with the magnetsand pole pieces fixed to the hub with adhesive, prior to theaddition of the containment sleeve. The axial segmentation ofthe magnets and pole pieces is a result of limitations in themanufacturing process.

V. EXPERIMENTAL RESULTS

Rotor version 3, shown in Fig. 14, was validated experimen-tally both by considering the electrical performance in relationto the baseline motor and by implementing an injection-basedsensorless position estimation strategy using the new machinedesign.

A. Electrical Performance

An experimental test rig, designed to derive the saliency ratioover a complete electrical or mechanical cycle, was used tocharacterize both the baseline machine and the new machineincorporating rotor version 3. The resulting saliency ratios forboth machines, averaged over a complete mechanical cycle,are shown in Table II. Also shown in Table II are the averagesaliency ratios calculated for both machines using FE modelingtechniques. Both the calculated and measured saliency ratiosare derived from the flux and current loci, respectively, resultingfrom the high-frequency injection signal at every rotor position[28]. These results highlight the good agreement between thetheoretical and measured values and show that, in comparisonto the baseline machine, the saliency of the machine incorpo-rating rotor version 3 is significantly increased as required forapplication of an effective sensorless control strategy.

TABLE IICALCULATED AND MEASURED AVERAGE SALIENCY RATIOS

Fig. 15. Normalized line-to-line back EMF. (a) Surface-mounted PM baselinemachine. (b) IPM rotor version 3.

The normalized back-EMF waveforms for both machinevariants, calculated using FE modeling techniques and mea-sured experimentally over one complete electrical cycle, areshown in Fig. 15. Fig. 15 highlights the good agreement be-tween the theoretical and measured back-EMF characteristicsand demonstrates the accuracy of the FE modeling techniqueemployed. The figure also highlights the agreement between thecharacteristics exhibited by the two machines. Since the rotortopologies are very different, the back-EMF waveforms are notexpected to be identical, and this is apparent from Fig. 15,


Fig. 16. Experimental sensorless control test platform.

Fig. 17. Uncompensated sensorless position.

which shows the different higher order harmonic content ineach waveform. However, the fundamental back-EMF compo-nents are very similar and suggest that the torque capability ofthe new rotor, which was a major design consideration, closelymatches that of the baseline machine.

B. Sensorless Position Estimation

An experimental test platform was used to verify the op-eration of a high-frequency injection-based sensorless rotorposition estimation scheme applied to the IPM rotor machine.

The test platform is based on a Texas InstrumentsTMS320C6713 digital signal processor (DSP) combined witha Xilinx XC2S300E field-programmable gate array (FPGA).The DSP generates the injection signals and processes thecurrents, measured by two Hall-effect current transducers andcaptured using analog-to-digital converters, in order to estimatethe rotor position. The FPGA is used to perform the pulsewidthmodulation (PWM) generation based on instructions from theDSP controller and to provide an interface between the DSP andthe analog I/O. A Mitsubishi intelligent power module is usedto provide power to the machine, and a shaft-mounted resolveris included to allow the accuracy of the estimation algorithmto be quantified. The test rig allows the easily interchangeablePM test motors to be loaded both statically and dynamically.Fig. 16 shows a block diagram of the experimental system witha locking gearbox for static testing.

With the machine operating under sensored control, theestimated rotor position, obtained using the high-frequencyinjection sensorless scheme where voltage is injected with anamplitude of 3 V and at a frequency of 300 Hz, and fundamentalcurrent Iq = 20 A, is shown in Fig. 17 and is compared withmeasured rotor position obtained from a resolver. It is important

Fig. 18. Saliency ratio versus per-unit torque for rotor version 3.

to note that the estimated rotor position is not compensated toaccount for distortion of the high-frequency injected voltage setintroduced by the PWM process [35].

The sensorless position signal shows distortion due to cog-ging but is in good agreement with the measured positionsignal. The results shown in Fig. 17 therefore confirm that themachine incorporating rotor version 3, which has been con-strained to meet existing mechanical and electrical performancecriteria, has a saliency profile that is suited to sensorless control.

VI. SALIENCY VARIATION

Magnetic saliency is highly dependent upon the operatingconditions of the machine. For example, high current may drivethe machine into saturation, or current may be phase advancedin order to optimize torque production, with the result thatthe degree of saliency will vary. This, in turn, will affect theaccuracy of saliency-dependent injection-based sensorless rotorposition estimation techniques.

Fig. 18 shows a simulation of the dependence of saliencyupon per-unit torque, at a fixed current angle, for the machineincorporating rotor version 3. The figure highlights the signifi-cant reduction in saliency ratio as torque (current) is increasedand shows that, above 1-p.u. torque, the saliency ratio reverses,i.e., Lq < Ld.

As stated in Section II, the machine is designed to operatewith a high peak-to-average torque ratio. Fig. 18 shows that


the machine would be capable of operating under sensorlesscontrol under load conditions up to approximately 0.5-p.u. peaktorque, thereby meeting the required operating specification. Afull treatment of the effects of saturation on the operation andperformance of the position-sensorless-controlled concentratedwound IPM machine will be presented in a forthcomingpublication.

VII. CONCLUSION

A high-frequency injection sensorless position estimationalgorithm has been directly incorporated into the FE design pro-cedure for an IPM machine, resulting in a machine that exhibitsthe required electrical characteristics while being tailored forsensorless control. To simplify presentation of the procedure,the analysis is based on the no-load condition. However, themethod could easily be extended to incorporate a particular loadcondition so that factors such as cross-coupling and saturationare captured in the analysis. This is the subject of ongoingresearch. The design was highly constrained such that the rotorwas required to fit within an existing stator and to produce thesame torque as the original motor. A simple manufacture andassembly process, which is compatible with existing aerospacepractices, was achieved using SMC pole pieces and a non-magnetic hub in place of the more usual laminated structure.The experimental results verify the machine performance andoperation under the control of an injection-based sensorlessrotor position strategy.

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Rafal Wrobel received the M.Sc.Eng. degree fromthe Technical University of Opole, Opole, Poland,in 1998 and the Ph.D. degree from the TechnicalUniversity of Lodz, Lodz, Poland, in 2000.

During 2001, he was an Assistant Professor withthe Technical University of Opole. Since 2002, hehas been a Research Fellow with the Universityof Bristol, Bristol, U.K. His research interests in-clude electromagnetic, thermal, and mechanical fieldanalysis and the optimization of electrical drives,with emphasis on permanent-magnet machines.

Alan S. Budden received the M.Eng. and Ph.D.degrees in electrical and electronic engineering fromthe University of Bristol, Bristol, U.K., in 2000 and2005, respectively. His Ph.D. research focused onsensorless rotor position measurement in permanent-magnet brushless machines at zero and low speeds.

He continued this work in collaboration withGoodrich Corporation, Actuation Systems, beforejoining the Controls Division, Rotork Controls Ltd.,Bath, U.K., where he is currently the Senior Elec-tronics Design Engineer that is responsible for new

technology research for industrial actuation.

Dan Salt received the M.Eng. degree in electricaland electronic engineering from the University ofBristol, Bristol, U.K., in 2007, where he is currentlyworking toward the Ph.D. degree in electrical andelectronic engineering.

His research interests include power electronics,high-performance ac machine drives, and sensorlesscontrol of permanent-magnet machines.

Derrick Holliday received the B.Eng. and Ph.D.degrees in electrical and electronic engineering fromHeriot-Watt University, Edinburgh, U.K.

After working as a Research Associate withHeriot-Watt University, he took a lectureship at theUniversity of Bristol, Bristol, U.K., where, as aSenior Lecturer, his research interests include high-efficiency electrical machines and drives, pulsewidthmodulation, sensorless control, and microelectro-mechanical systems.

Phil H. Mellor received the B.Eng. and Ph.D. de-grees in electrical engineering from the Universityof Liverpool, Liverpool, U.K., in 1978 and 1981,respectively.

He held academic positions at the University ofLiverpool from 1986 to 1990 and The University ofSheffield, Sheffield, U.K., from 1990 to 2000. He iscurrently a Professor of electrical engineering withthe University of Bristol, Bristol, U.K. His researchactivities include high-efficiency electric drives andactuation and generation systems for application in

more electric aircraft and hybrid electric vehicles.

Andrei Dinu received the Ph.D. degree in electricaland electronic engineering from De Montfort Uni-versity, Leicester, U.K., in 2000, with a thesis con-cerning the sensorless control of induction motorsusing hardware-implemented neural networks.

In August 2000, he was appointed Lecturer atDe Montfort University, where he conducted re-search in the field of electrical drive control until2003, when he moved to industry. He first workedas a Design Engineer with Datalink Electronics,Loughborough, U.K. In 2004, he joined Goodrich

Corporation, Birmingham, U.K., as a Control Systems Engineer, where he iscurrently an Engineering Consultant involved in R&D with the ElectromagneticSystems Technical Center.

Parmider Sangha received the B.Sc. degree inelectrical engineering from Newcastle University,Newcastle upon Tyne, U.K., and the Ph.D. degreefrom the University of Bath, Bath, U.K.

He is currently an Engineering Consultant withElectrical Power Systems, Goodrich Corporation,Electrical Power Systems, Birmingham, U.K. Hismain interests include electromagnetics, the devel-opment of numerical analysis techniques, and thedesign and development of electrical machines foraerospace applications.

Mark Holme received the Honours degree in engi-neering from Coventry Polytechnic, Coventry, U.K.,and the Masters degree in power electronics anddrives from the University of Birmingham, Birming-ham, U.K.

He has over 20 years of experience in the fieldof power electronics and motor drives, covering re-search and development, product design, and projectmanagement, and has led a number of programs tofurther the application of motor drives in aircraft ap-plications. He is currently the Engineering Manager

of motor drive systems with Engine Control and Electrical Power Systems,Goodrich Corporation, Birmingham.

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Sensorless Position PIM