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Korean Instituteof ElectromagneticEngineering Society
AWAP 2016Asian Workshop on Antennas and Propagation
http://www.kiees.or.kr/awap2016
January 27-29, 2016, Centum Hotel, Busan, Korea
Organized by
Sponsored by
Korean Instituteof ElectromagneticEngineering Society
Korea Advanced Institute of Science & Technology
Technical Group on Antennas and Propagation of the Korean Institute of Electromagnetic Engineering and Science
Technical Committee on Antennas and Propagation of the Institute of Electronics, Information and Communication Engineers
Electromagnetic Group of the Electrical Engineering/Electronics, Computer, Telecommunications and Information Technology Association
IEEE Antennas and Propagation Society, Seoul Chapter
- i -
Message from General Chairmen ........................................................ ii
Organizing Committee .......................................................................... iii
PROGRAM .............................................................................................. iv
Regular Session 27th January, 2016, Wednesday .................................................1
Plenary Talk and Invited Session 28th January, 2016, Thursday .......................23
Map ........................................................................................................47
Banquet .................................................................................................48
Transportation ........................................................................................49
Contents
- ii -
Welcome to AWAP2016 in Busan!
It is a great pleasure and a distinct honor to host the 3th 2016 Asian Workshop on Antennas and Propagation
(AWAP2016) which will be held at Centum Hotel in Busan, Korea, on January 27-29, 2016.
It has been a pleasure to work with our colleagues who have organized the program. Prof. Jae-Young Chung,
Prof. Minseok Kim, and Prof. Akkarat Boonpoonga helped put together a very strong and technical program
for comprising 34 papers. We do hope that these papers will be found to be intriguing and of enhanced scientific
quality by the participants. In this workshop, it will be organized with the single oral session delivered by invited
researchers and professors to encourage intimate professional discussion and the two parallel sessions for the
award contest.
AWAP2016 is a continuation of a series of annual antenna workshops held in Kanazawa, Japan (2014) and
Bangkok, Thailand (2015). The AWAP2016 is an annual forum for the exchange of information on the research
and development in antennas technologies, propagation, and related fields. AWAP2014 was the first AWAP and
was grown from the joint conference KJAP of two countries to the Asian Workshop in which all researchers and
students were welcomed from any countries from all over the world. The 2014 Asian Workshop on Antennas and
Propagation was held at the Kanazawa Theatre in Kanazawa, Japan from May 14 to 16, 2014. The 2015 Asian
Workshop on Antennas and Propagation (AWAP2015) was held at Swissotel Le Concorde in Bangkok, Thailand,
from June 17 (Wednesday) to 18 (Thursday), 2015.
The AWAP2016 is jointly organized by the Technical Group on Antennas and Propagation society of the Korean
Institute of Electromagnetic Engineering and Science (KIEES), Korea, the Technical Committee on Antennas
and Propagation of the Institute of Electronics, Information and Communication Engineers (IEICE), Japan, and
Electromagnetic Group of the Electrical Engineering/Electronics, Computer, Telecommunications and Information
Technology Association (ECTI), Thailand.
We would like to express our sincere gratitude to every author and participant whose high-level contributions
guaranteed the success of the AWAP2016. Moreover, we warmly thank Prof. Kyeong-Sik Min and the members
of the International Steering Committee as well as the Technical Program Committee for their constant help and
constructive advice.
Finally, thanks to the members of the Organizing Committee and the Secretariat for their support in the
completion of the Conference.
We hope that AWAP2016 will be stimulating, enjoyable, and fulfilling experience to all who attend it.
Wishing all the members a happy and prosperous 2016!
General Co-Chair of AWAP2016
Professor Seong-Ook Park (Korea)
Professor Qiang Chen (Japan)
Professor Titipong Lertwiriyaprapa (Thailand)
Message from General Chairmen of AWAP2016
- iii -
International Advisory Committee Members
Prof. Toshikazu Hori (University of Fukui, Japan)
Prof. Hiroyuki Arai (Yokohama National University, Japan)
Prof. Jaehoon Choi (Hanyang University, Korea)
Prof. Kin-Lu Wong (National Sun Yat-Sen University, Kaohsiung, Taiwan)
Prof. Monai Krairiksh (King Mongkut’s Institute of Technology Ladkrabang, Thailand)
Prof. Young Joong Yoon (Yonsei University, Korea)
Prof. Sangwook Nam (Seoul National University, Korea)
Prof. Kyeong-Sik Min (Korea Maritime and Ocean University, Korea)
Dr. Jin-Seob Kang (KRISS, Korea)
Prof. Keizo Cho (Chiba Institute of Technology, Japan)
Conference Co-Chairs
Prof. Seong-Ook Park (Korea Advanced Institute of Science and Technology, Korea)
Prof. Qiang Chen (Tohoku University, Japan)
Prof. Titipong Lertwiriyaprapa (King Mongkut's University of Technology, North Bangkok, Thailand)
Technical Program Committee Co-Chairs
Prof. Kangwook Kim (Gwangju Institute of Science and Technology, Korea)
Dr. Masayuki Nakano (KDDI R&D Laboratories Inc.)
Prof. Sarawuth Chaimool (Udon Thani Rajabhat University, Thailand)
Member: Prof. Kyung-Young Jung (Hanyang University, Korea)
Member: Prof. Keum Cheol Hwang (Sungkyunkwan University, Korea)
Member: Prof. Ick-Jae Yoon (Chungnam National University, Korea)
Local Arrangement
Prof. Dong-Kook Park (Korea Maritime and Ocean University, Korea)
Prof. Joong Han Yoon (Silla University, Korea)
Secretaries
Prof. Jae-Young Chung (Seoul Nat’l University of Science and Technology, Korea)
Prof. Minseok Kim (Niigata University, Japan)
Prof. Akkarat Boonpoonga, (King Mongkut's University of Techonolgy, North Bangkok, Thailand)
Organizing Committee
- iv -
PROGRAM
Number Start Min. 27th January, 2016, Wednesday, Regular Session
Session-R1 Session-R2
Authors Affilliation Title Chair Room Authors Title Affilliation Chair Room
1 15:00 15
Minseok Kim*, Tatsuki Iwata, Kento Umeki, Karma Wang-chuk, Jun-ichi Takada, and Shigenobu Sasaki
Niigata University, Tokyo In-stitute of Technology
Identification of Propagation Mechanism of Mm-Wave Outdoor Access Link
Keum Cheol Hwang
831(4th floor)
Nu Pham and Jae-Young Chung*
A dual-band GPS an-tenna integrated inside a military helmet
Seoul Nat'l Univer-sity of Science and Technology
Kyung-Young Jung
832(4th floor)
2 15:15 15Haewon Jung and Kangwook Kim*
Gwangju Institute of Science and Tech-nology
Subsurface GPR Imaging of Pavement Using MULSM Method
Minseok Kim
Naobumi Mich-ishita*, Naoto Nishiyama, Hisashi Morishita
Helmet Folded Dipole Antennas
National Defense Academy, Japan
Jae-Young Chung
3 15:30 15
Lakkhana Ban-nawat, Feaveya Kheawprae, and Akkarat Boonpoonga*
King Mongkut's University of Technol-ogy North Bankok
Improvement of Radar Target Identification with Near-field Calibra-tion Technique
Kang-wook Kim
Kunio Sakakibara*, Kei Firdaus, No-buyoshi Kikuma
Design of Frequency Selective Spiral Slot located in Near-Field of Wireless Power Trans-fer System by Eigen Mode Analysis
Nagoya Institute of Technology
Naobumi Michishita
4 15:45 15Jun Gi Jeong* and Young Joong Yoon
Yonsei Uni-versity
Gain enhanced compact bow-tie antenna with director
Akkarat Boon-poonga
Do-Gu Kang* and Jaehoon Choi
PIFA antenna for UWB applications
Hanyang Univer-sity
Kunio Sakaki-bara
16:00 15 Break
5 16:15 15Takeshi Fuku-sako*, Shohei Higashi
Kumamoto University
A Sensor Antenna for Non-destruc-tive Testing
Jae-Young Chung
831(4th floor)
Sarawuth Chaimool*, Prayoot Akkaraekthalin, Kwok L. Chung
Wideband Sequential-rotation arrays with circularly polarized Patch radiators using Anisotropic Metasur-face
Udon Thani Ra-jabhat University, King Mongkut's University of Technology North Bankok, Qingdao Technological Uni-versity
Seong-Ook Park
832(4th floor)
6 16:30 15Bo-Hee Choi and Jeong-Hae Lee*
Hongik University
4x4 Loop array for magnetic field control of wireless power transfer
Takeshi Fuku-sako
Myeongjun Kong, Geonyeong Shin, and Ick-Jae Yoon*
Electrically small spherical antennas using 3D printing tech-nology
Chungnam Nat'l University
Sarawuth Chaimool
7 16:45 15
Yuichi Kimura*, Fumihiko Nonaka, and Sakuyoshi Saito
Saitama University
Standing-wave and traveling-wave excitation of a microstrip antenna array fed by transverse slots on a broad wall of the rectangular waveguide for linear polarization parallel to the axis
Jeong-Hae Lee
Soon-Soo Oh*, Dong-Woo Kim, Tae-Hyung Kim, and Chi-Hyung Ahn
Beamwidth Reconfigu-rable Array Antenna Without Power Loss Using the Switched Coupler
Chosun University, Korea Railroad Research Institute
Ick-Jae Yoon
8 17:00 15
Jang-soon Park*, Jun-Bong Ko, and Dongho Kim
Sejong University
A Frequency-Reconfigurable Dipole Antenna Using a Tapered Impedance Match-ing Structure
Yuichi Kimura
Byeong-Yong Park, Tae-Wan Kim, and Seong-Ook Park*
Analysis of mode Splitting Behavior for Cylindrical Ferrite Resonator Antenna
Korean Advanced Institute of Science and Technology
Soon-Soo Oh
17:15 45 Break
18:00 150Welcome Reception
20:30
- v -
number Start Min. 28th January, 2016, Thursday, Plenary Talk and Invited Session
8:30 40 Registration
9:10 15 Opening Ceremony, 831(4th floor)
9:25 30Plenary Talk, 831(4th floor) Chair
Toshikazu Hori* University of Fukui Low-Profile Design of Meta-Surface with Frequency Selective Surface and Its Application
Seong-Ook Park
Invited Session, 831(4th floor)
Authors Affilliation Title Chair
1 9:55 20 C. Rienthong, C. Kittiyanpunya, and M.Krairiksh*
King Mongkut’s Institute of Technology Ladkrabang
Surface moisture content sensor detecting mutual coupling magnitude between parallel and perpen-dicular dipole antennas (Invited paper)
Toshikazu Hori
2 10:15 20 Jinpil Tak, Eun Jeong, and Jaehoon Choi* Hanyang University Design of a Metamaterial Absorber for 24 GHz Auto-
motive Radar System (Invited paper)Monai Krai-riksh
10:35 15 Break
3 10:50 20 Ikmo Park* and Son Xuat Ta Ajou University Cavity-Backed Printed-Dipole Antenna for Millime-ter-Wave Applications (Invited paper) Jaehoon Choi
4 11:10 20 Jiro Hirokawa*, Dong-Hun Kim Tokyo Institute of Technology Waveguide Short-slot 2D-plane Coupler for 2D Beam-switching Butler Matrix (Invited paper) Ikmo Park
5 11:30 20 Ji Hwan Yoon and Young Joong Yoon* Yonsei University Millimeter-wave Reflectarray Antennas with Dual-reflector Configurations (Invited paper) Jiro Hirokawa
6 11:50 20 Yoshio Inasawa*, Takashi Tomura, Michio Takikawa, Hiroaki Miyashita
Mitsubishi Electric Corpora-tion
Gain Improvement of Shaped-beam Reflector Using Simultaneous Design of a Multimode Horn and Shap-ing Functions (Invited paper)
Young Joong Yoon
12:10 90 Lunch, Steering Committee meeting (18th floor meeting room)
7 13:40 20 Kin-Lu Wong* National Sun Yat-sen Univer-sity
Introduction to 5G Communications and its Smart-phone Antenna Design Perspectives (Invited paper) Yoshio Inasawa
8 14:00 20 Seungtae Ko*, Youngju Lee, Kwanghyun Baek, Yoongun Kim and Wonbin Hong Samsung Electronics Low Profile PCB Integrated mmWave Array Antenna
Solutions for 5G Mobile Communication (Invited paper) Kin-Lu Wong
9 14:20 20 Kentaro Nishimori* Niigata University Multi-beam massive MIMO using analog-digital hybrid configuration (Invited paper)
Jae-Young Chung
14:40 15 Break
10 14:55 20Yuya Tojima, Hiroki Sudo, Takayuki Kubota, Keizo Cho*, Hiroaki Nakabayas-hi, Koji Suizu
Chiba Institute of Technology Measurement of Antenna Substrate by Collimated THz Waves (Invited paper)
Kentaro Nishi-mori
11 15:15 20 Kyeong-Sik Min* Korea Maritime and Ocean University
High-gain Multiband Spiral Antenna Design History for NLJD System (Invited paper) Keizo Cho
12 15:35 20 T. Imai*, K. Kitao, N. Tran, N. Omaki, Y. Okumura, and K. Nishimori
NTT DOCOMO, Niigata Uni-versity
A Study on Penetration Loss Modeling for 0.8 to 37 GHz Band (Invited paper)
Kyeong-Sik Min
13 15:55 20 Jin-Seob Kang*, Jeong-Hwan Kim, and Jeong-Il Park
Korea Research Institute of Standards and Science
Parameter Comparison of Standard Gain Horn Antenna at R-/S-/X-Band (Invited paper) Tetsuro Imai
16:15 15 Break
14 16:30 20 Titipong Lertwiriyaprapa* and Montree Saowadee
King Mongkut’s University of Technology North Bangkok, Anunda Technology
Development of an Approximate UTD Ray Solution for EM Diffraction by a Planar Material Junction on PEC Ground Plane (Invited paper)
Jin-Seob Kang
15 16:50 20 Il-Suk Ko* Inha UniversityDirect Derivation of Closed-form Expression of Sommerfeld Integral for Impedance Half-plane from Exact Image Formulation (Invited paper)
Titipong Ler-twiriyaprapa
16 17:10 20 Hiroyuki Arai* Yokohama National Univer-sity
Optical beam scanning antenna for ultra high speed short range communication system (Invited paper) Il-Suk Ko
17 17:30 20 Sangwook Nam* Seoul National University An Electrically Small Isotropic Antenna Using Folded Split Ring Resonator (Invited paper) Hiroyuki Arai
17:50 10 Break
18:00 180Banquet
21:00
Start Min. 29th January, 2016, Friday, Technical Tour and Discussion
9:00 180Technical Tour and Discussion
12:00
- 1 -
AWAP 2016Asian Workshop on Antennas and Propagation
27th January, 2016Wednesday
Regular Session
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Identification of Mm-Wave Radio Propagation Mechanism inOutdoor Access Links
Minseok Kim†, Tatsuki Iwata†, Kento Umeki†, Karma Wangchuk‡, Jun-ichi Takada‡, Shigenobu Sasaki†,†Graduate School of Science and Technology, Niigata University, Niigata, Japan
‡Graduate School of Science and Engineering, Tokyo Institute of Technology, Tokyo, JapanEmail: [email protected]
Abstract—This paper discusses the dominant propagationmechanisms in an outdoor environment through the channelmeasurement and ray tracing simulation at a mm-wave band of58.5 GHz where the measurements were conducted in an outdoorenvironment in Niigata university campus assuming the open areaoutdoor hotspot access scenario in 5G mobile systems.
I. INTRODUCTION
In future mobile systems, it will be necessary to operatein densely populated areas with increasing capacity and datarates to a much greater degree, and hence conventional cellularnetworks covering as large area as possible cannot be expectedto provide sufficient performance any longer. To develop superhigh bit-rate systems beyond 4G, there is a general consensusthat signal bandwidth should be significantly increased inaddition to utilize recent powerful transmission techniquessuch as MIMO (multiple-input-multiple-output technology),coordinated multipoint (CoMP), heterogeneous networks (Het-Nets), and carrier aggregation (CA). However, because ofserious congestion of the frequency spectrum of lower mi-crowave bands below 6 GHz, developing new frequency bandsshould be inevitable choices. Obviously, it is well knownthat free space propagation loss and shadowing loss are bothsignificantly increasing with frequency increase, which limitscommunication range. That is a reason why a small cellcommunication using high frequency bands within a confinedarea is currently gaining much attention [1], [2].
Currently, lots of studies argue the use of microwave andmillimeter wave (mm-wave) spectrum for cellular networkssuch as 28 and 38 GHz [1] which are allocated for localmultipoint distribution service (LMDS) and currently availablewith spectrum allocations of over 1 GHz bandwidth as wellas 60 GHz which offers 5 ∼ 9 GHz of unlicensed bandwidthin most countries [2]. However, radio propagation channelproperties at high frequency bands in such small cell mobileapplications have not been sufficiently studied. Moreover,there are few reports on the propagation properties in outdoorenvironments and the dominant propagation mechanisms havenot been thoroughly investigated by measurements.
In small cell environments, site-specific property of thepropagation mechanism is very important to develop a betterchannel model. As an initial step, in this study, the mm-wave propagation mechanism in an outdoor environment isidentified through the channel measurement and ray tracing
BS
MS
Fig. 1. Measurement campaign (topview).
(RT) simulation at a mm-wave band of 58.5 GHz where themeasurements were conducted in an outdoor environment inNiigata university campus assuming the open area outdoorhotspot access scenario in 5G mobile systems.
II. MEASUREMENT CAMPAIGN
In the measurement, the developed costom channelsounder has been used, which employs a commercialproduct of mm-wave Tx and Rx which integrate waveg-uide module with standard WR15/WG25 flange interfaces(V60TXWG1/V60RXWG1, VubIQ) [3]. It is configured in2 × 2 MIMO to measure full polarimetric channel responsesimultaneously. The RF transceivers employ a heterodyneIF architecture with variable frequency IF and RF mixersfor different RF channel selection, which requires a singlecommon synthesizer for IF and RF LO signal generation. Inthe typical setup, the baseband signal input power is adjustedby approximately −13 dBm, so that the transmit power ofapproximately 10 dBm is achieved by the power amplificationof 23 dB. We exclude the influence of the measurement systemfrom the measured channel responses by full MIMO back-to-back calibration (direct connection between Tx and Rx antennaports with a waveguide and an attenuator). The measurementdynamic range is limited to approximately less than 40 dB.
The measurement campaign was conducted in an outdooropen area as shown in Fig. 1 where Tx which was assumedto be the base station (BS) was located at around the centerof the area and the channel transfer functions were measuredat three MS positions. MS pos1 and MS pos2 were in line-of-sight (LoS) condition and MS pos3 was in obstructed-LoS(OLoS) condition. The antenna heights were 3 m for BS and1.5 m for MS. The area is surrounded by some buildings which
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The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
(a) MS pos1 (VV) (b) MS pos2 (VV)
(c) MS pos3 (VV) (d) MS pos3 (HH)
Fig. 2. Synthetic PDPs; (a)(b):LoS, and (c)(d):OLoS.
is 20 ∼ 30m far away from the BS antenna. The distancebetween BS and MS antennas was approximately 30 m.
For directional channel acquisition, high gain horn antennaswere rotated. The gains and half power beam-width (HPBW)are 15 dBi and 30 degrees for MS, and 24 dBi and 12 degreesfor BS, respectively. The BS and MS antenna were rotatedfrom 0 to 360 degrees in azimuth, and from 60 to 120 degreesin co-elevation. Azimuth and co-elevation at BS and MS werevaried in 12 and 30 degree steps, respectively.
III. RESULTS
From the measurement data, we obtained double directionalangle delay power spectrum (DDADPS). Then, the angularpower spectrum (APS) at both sides of the BS and MS,and the omni-directional power delay profile (PDP) weresynthesized from the DDADPS. For precise interpretation ofthe measurement results, the RT simulation was used wherethe maximum order of reflection were set to be three for LoSand two for OLoS conditions, respectively. The RT simulationemploys the image method. The first order diffraction wasfurther calculated only for OLoS condition based on uniformtheory of diffraction (UTD). This simulation calculated theray parameters of the received power, time delay of arrival,angles of departure and arrival for each path. For comparisonwith the measurement results, the simulation based DDADPSwas reconstructed from those parameters using the antennapatterns, then the APS and PDP were calculated in the samemanner as the measurement.
From the synthetic omni-directional PDPs of Fig.2, it canbe seen that a few significant multi-paths are observed besidesLoS path in the limited measurement dynamic range, and thedominant paths in the RT results are well matched to thosein the measurement results. On the other hand, using themeasured and simulated APS at both sides of BS and MS
(a) APS@BS
(b) APS@MS
Fig. 3. Synthetic APSs at 107.5 ns for MS pos3 (OLoS condition).
for the individual delay tap, the propagation mechanism wasidentified. The APS at delay tap of 107.5 ns for MS pos3 areshown in Figs.3. It illustrates that the propagation mechanismof that path is the edge diffraction on the vertical metallicpillar, which is supported by the PDP of MS pos3 showingHH-pol has a larger gain than VV-pol in Figs.2(c) and (d). Inthe same manner, all dominant propagation mechanisms wereidentified up to the third order specular reflection from wallsand the corresponding ground reflection, the penetration intoglass, and the first order diffraction. Some other observationsare summarized as follows.
• Only a few significant multi-paths were observed. Thesecond largest path powers for MS pos1 and MS pos2were 7 dB and 15 dB below LoS power, respectively.
• In OLoS condition, the path gain in HH-pol is sig-nificantly larger than VV-pol where the power of thediffracted path on the vertical edge was less than 5 dBfrom that of the largest reflected path in HH-pol.
• The detected paths include the corresponding groundreflected path due to the low measurement resolution. Theground reflection should be appropriately considered inthe channel model as a shadowing factor.
ACKNOWLEDGMENT
This work was partly supported by “The Strategic In-formation and Communications R&D Promotion Program(SCOPE: No.145004102)” and JSPS KAKENHI Grant Num-ber 15H04003.
REFERENCES
[1] T. Rappaport, et al., “Millimeter Wave Mobile Communications for 5GCellular: It Will Work!,” IEEE Access, Vol. 1, 2013.
[2] MiWEBA, FP7 ICT-2013-EU-Japan, http://www.miweba.eu[3] M. Kim, K. Umeki, K. Wangchuk, J. Takada, S. Sasaki, “Polarimet-
ric Mm-Wave Channel Measurement and Characterization in a SmallOffice,” Proc. PIMRC 2015, Aug. 2015.
4
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Subsurface GPR Imaging of Pavement Using MULSM Methodo Haewon Jung and Kangwook Kim
Gwangju Institute of Science and Technology
[email protected] and [email protected]
Ⅰ. IntroductionIn pavement inspection, the ground penetrating radar
(GPR) is frequently used to inspect the subsurface
structures, such as pavement thickness, reinforcement bars
(rebars), water pipes, etc. It can also be used to detect
subsurface cavities nondestructively with high resolution
features.
Generally, the pavement is a multilayer geometry with
asphalt, concrete, sand, and etc. In a multilayer geometry,
multilayer Stolt migration (MULSM) method can be used
to produce migrated images [1]. The MULSM is a hybrid
method in such a way that the phase-shift migration [2] is
recursively applied to each boundary between layers, and
then the Stolt migration [3] is used within the
homogeneous layer under each boundary.
In this work, GPR survey was conducted in experimental
pavement geometry, and the collected data was imaged to
identify subsurface targets using the MULSM method.
Ⅱ. Experiment and ImagingThe pavement geometry that contains cavities, brick,
metal sheets, and rebar is illustrated in Fig. 1, where the
cavities are modeled as a block of Styrofoam. The top and
bottom layers are asphalt and sand with relative
permittivity of approximately 6 and 4, respectively.
The antenna array data is obtained from a synthesized
aperture of monostatic radar that is composed of a vector
network analyzer, resistive vee dipole antenna [4]. The
input waveform is a differentiated Gaussian pulse with a
peak frequency at 2.25 GHz.
The migrated result under pavement is depicted in Fig. 2
as an isosurface image. The targets are seen to be well
migrated under multilayer pavement geometry.
Ⅲ. ConclusionIn this work, the GPR is used to collect data under
pavement that contains objects. The data is migrated using
MULSM method and the results are well matched to the
actual target positions.
Figure 1. Experiment setup.
Figure 2. Isosurface image of migrated result.
Acknowledgment
This work was supported by ICT R&D program of
MSIP/IITP [10041950, Development of mobile safety-
inspection systems using high resolution penetration
imaging technology for transportation infrastructure].
References[1] Y. C. Kim, R. Gonzalez, and J. R. Berryhill,
“Recursive wavenumber-frequency migration,” Geophysics, vol. 54, no. 3, pp. 319-329, Mar. 1989.
[2] J. Gazdag, “Wave equation migration with the phase-shift method,” Geophysics, vol. 43, no. 7, pp. 1342-1351, Dec. 1978.
[3] R. H. Stolt, “Migration by Fourier transform,” Geophysics, vol. 43, no. 1, pp. 23-48, Feb. 1978.
[4] K. Kim and W. R. Scott, “Design of a resistively loaded vee dipole for ultrawide-band ground-penetrating radar applications,” IEEE Trans. Antennas Propagat., vol. 53, no. 8, pp. 2525-2532, Aug. 2005.
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The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Improvement of Radar Target Identification
with Near-field Calibration Technique Lakkhana Bannawat, Feaveya Kheawprae, and Akkarat Boonpoonga
Department of Electrical and Computer Engineering, Faculty of Engineering, King Mongkut’s University of Technology North Bangkok, Thailand
[email protected], [email protected] and [email protected]
Ⅰ. Introduction Radar target identification has been extensively studied
in research area of electromagnetic wave propagation.
After transmitting an impulse signal to a target,
electromagnetic field scattered from the target is utilized to
detect and characterize objects of various shapes and
constitutions. The singularity expansion method (SEM)
was introduced to model the late- time portion of the
electromagnetic transient response of a target irradiated by
an electromagnetic pulse as a sum of damped exponentials
with complex natural frequency [1]. A complex frequency
which is often referred as a pole extracted from late time
response is a tool for aspect-independent target
identification. A Matrix pencil method (MPM) is one of the
most popular techniques which are widely employed to
extract the poles [2]. Recently the MPM was slightly
modified as the short-time matrix pencil method (STMPM)
which can resolve the problem of finding the
commencement of late-time portion [3].
In the paper, we show another problem of applying the
SEM to identify the object in the practical situation. In the
radar system, the antenna is an essential component to
transmit and receive the EM pulse. However, this
component impact on the accuracy of pole. This paper
presents a calibration technique to reduce the degradation
of pole due to the antenna response.
Ⅱ. Simulations and Results Simulations are conducted to verity the proposed
technique by using electromagnetic software simulator.
The PEC cube with dimension of 20 cm and sphere with
radius of 20 cm are modeled as radar targets. The bow-tie
antenna is employed as both transmitting and receiving
antennas. The distance between the antennas is 60 cm. The
calibration technique proposed in [4] is applied to reduce
the effect of the antenna response on the pole extracted by
using STMPM. Figure 1 and 2 show the natural frequency
extracted by using STMPM without and with the proposed
technique, respectively. At the late-time portion, note that
the natural frequencies of PEC sphere and cube, extracted
by using STMPM without the calibration technique are
almost identical. These frequencies cannot be used to
identify the targets. To resolve this problem, the calibration
technique proposed in [4] is applied before extracting poles.
Figure 2 clearly reveals that the nature frequencies of PEC
sphere and cube, extracted by using STMPM without the
calibration technique is separate.
Figure 1. Natural frequency extracted by STMPM without
the proposed technique.
Figure 2. Natural frequency extracted by STMPM with the
proposed technique.
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The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Ⅲ. Conclusion This paper has presented an improvement of a radar
target identification by using a near-field calibration
technique. In the radar system, a target can be identified
based on a singularity expansion method (SEM) principle.
Poles representing the signature of a target is extracted by
using a short-time matrix pencil method (STMP).
Transmitting and receiving antennas are the important parts
of the radar system for transmitting and receiving the radar
pulse. In the practical situation, the response of the antenna
impacts on the accuracy of extracted poles. To resolve the
underlying problem, we introduce an efficient technique of
the antenna calibration technique in order to reduce its
effect. Simulations were conducted to verify the proposed
technique. The results show the increase of accuracy of
extracted pole.
References [1] C. E. Baum, E. J. Rothwell, K. Chen, and D. P.
Nyquist, “The Singularity Expansion Method and Its Application to Target Identification,” Proceedings of the IEEE, Vol. 79, No. 10, Oct. 1991
[2] T. K. Sarkar and O. Pereira, “Using the matrix pencil method to estimate the parameters of a sum of
complex exponentials,” IEEE Antenna and Wireless propagation magazine, vol. 37, pp. 44-55, 1995.
[3] R. Rezaiesarlak and M. Manteghi, “Short-Time
Matrix Pencil Method for Chipless RFID Detection
Applications,” IEEE Trans. on Antennas and Propagation, Vol. 61, No. 5, May 2013.
[4] V. A. Mikhnev and P. Vainikainen, "Single-
Reference Near-Field Calibration Procedure for Step-
Frequency Ground Penetrating Radar," IEEE Trans.
on Geoscience and Remote Sensing, vol. 41, No. 1
pp. 75-80, Jan. 2003.
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The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Gain Enhanced Compact Bow-tie Slot Antenna with Director oJun Gi Jeong, Young Joong Yoon
Department of electrical and electronic engineering, Yonsei University
.Ⅰ IntroductionUltra-wide Band (UWB) applications are used in many
modern systems, and several types of antennas are used. In
many cases, bow-tie type antenna is used for the UWB
applications because of the dimensional compactness and
the broadside pattern. Typically, reflector is used at the
back side of antenna to make the unidirectional pattern. In
addition, several methods are proposed to enhance the gain
of bow-tie antenna[1] for high gain applications recently.
However, previous cases are occupy a large area.
In this paper, gain enhanced compact bow-tie antenna is
proposed. It has a director to enhance the gain of antenna
and also it has compact size. Thus, this characteristics are
useful for compact UWB systems for high gain.
.Ⅱ Design and resultDesigned antenna is shown in Fig. 1. The proposed
antenna has three layers and theses are designed in the
same planar area. Director is composed of short lines and it
is arrayed in both sides (top, bottom) of substrate.
The Electric field from the bow-tie antenna is induce the
current at the short lines of the director, and electric field is
re-radiated from the short lines. It is similar with the
operation principle of quasi Yagi-Uda antenna.
Designed antenna is operating at the upper band of
UWB (6.09 ~ 11.28 GHz) as shown in Fig. 2. Also,
antenna gain is enhanced about 0.5 ~ 1.6 dB at the overall
operating band as shown in Table 1.
Antenna parameters are determined as L=31mm,
W=13mm, h1=11mm, h2=7mm, l1=13mm, l2=4.2mm,
l3=6.35mm, respectively.
Figure 1. Configuration of proposed antenna.
Figure 2. S-parameter of proposed antenna.
Table 1. Comparison of antenna gain [dB]
7GHz 8GHz 9GHz 10GHz
Without director 6.2 7.8 8.5 8.5
With director 6.7 8.6 9.5 10.1
.Ⅲ ConclusionA gain enhanced compact bow-tie slot antenna is
proposed. It is operating at the upper band of UWB. Also,
the gain of antenna is enhanced in overall operation band.
Thus, the proposed antenna can be used to high gain UWB
applications.
Acknowledgement
This research was supported by the MSIP(Ministry of
Science, ICT and Future Planning), Korea, under the
ITRC(Information Technology Research Center) support
program(IITP-2015-H8501-15-1019) supervised by the
IITP(Institute for Information & communications
Technology Promotion)
References
[1] Shi-Wei Q., Chi-Hou C. and Quan X.,
“Ultrawideband composite cavity-backed folded
sectorial bowtie antenna with stable pattern and high
gain," IEEE Trans. Antennas and Propagation, vol.
57, no. 8, Aug. 2009.
8
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
A Sensor Antenna for Non-Destructive Testingo Takeshi Fukusako and Shohei Higashi, Kumamoto University
1. IntroductionSensor antennas have been widely used recently in several fields. A sensor antenna using RFID Chiphas been proposed for detecting the relative permittivity of material[1][2]. Although there is a variety of sensing method for concrete, we proposed a sensor antenna which can detect the relative permittivity of concrete by measuring the resonant frequency and transmitted power. 2 The sensing system
A sensor antenna and a reader antenna are used in the proposed sensing system. The sensor antennaequipped with RFID is pasted on a concrete block.The sensor antenna is provided with the power through the reader antenna. When concrete is changed from dry state to wet one, the relative permittivity of concrete becomes high. In this case, the resonant frequency of sensor antenna is shifted lower. The readercan detect the shifted frequency with scanning. The frequency shift indicates the wed degree of concrete.3 Antenna StructureFig.1 shows the proposed structure. The proposed antenna consists of four parts consisting of ground plane, feed part, meander-line antenna and additional microstrip elements. There is a slit in the ground plane in order to be sensitive for the permittivity shift of concrete. One tip of the meander line antenna is shorted to feed part. In addition, four microstrip elements along a slit nearby the shorting part. As shown in Fig. 2, this sensor antenna is pasted on the concrete, and a dipole antenna is used as the reader antenna. The distance between the sensor antenna and the reader antenna is 1 m.4 Simulation resultsFig.3 shows simulation results, where S11 and S21 characteristics are analyzed when the relative permittivity of the concrete is between 4 and 10.S11 characteristics are shifted low with an increase of the relative permittivity. In the S21 characteristics, the reader antenna can obtain high transmitted power with a keep peak at the resonant frequency when the relative permittivity of concrete is 4,5,8 and 10. However, the keen peak is not clear for other values of relative permittivity.5 ConclusionA sensor antenna for non-destractive testing has been proposed. The proposed antenna has a narrow band characteristics so as to make a keen peak in transmitted power characteristics. This contribute to a precise measurement of permittivity.
Fig.1 Proposed sensor antenna structure
References [1]R.Suwalak , C.Phongcharoenpanich , D.Torrungrueng, and M.Krairiksh,“DETERMINATION OF DIELECTRIC PROPERTY
OF CONSTRUCTION MATERIAL PRODUCTS USING A NOVEL RFID SENSOR " Progress In Electromagnetics Research , Vol.130, 601-617, 2012[2] F. Yang, Q. Qiao, and A. Z. Elsherbeni, “Reconfigurable sensing antennas: concept, design, and applications,” Antennas
and Propagation in Wireless Communication (APWC2013), pp. 748-752, Torino, Italy, September 2013.
Fig.2 Simulation model
Fig.3 Simulated results
9
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
4x4 Loop array for magnetic field control of wireless power transfer
Bo-Hee Choi and o Jeong-Hae Lee
Department of Electronic Information and Communication Engineering, Hongik University,
Seoul 121-791, Korea
Ⅰ. IntroductionAngular misalignment of a receiver is sensitive issue of
wireless power transfer (WPT) because it reduces power
transfer efficiency (PTE). Three-dimensional structure with
multi-sources was researched for magnetic-field control [1].
Later on, planar-type loop array controlling magnetic-field
with a source was presented [2]. It has an advantage for
practical space in usage and it can be a key technology to
improve a reduced PTE of a misaligned receiver. In this
paper, a 4x4 loop array for magnetic-field control will be
presented to improve a PTE of a misaligned receiver in
wireless power transfer.
Ⅱ. Magnetic-field control loop arrayFig. 1 shows the configuration and dimension of 4x4
loop array with a source at the first loop. A receiver is
50cm away from the center of array and can be rotated 0°,
45°, and 90°. The receiver size is the same as that of one
array element. The operating frequency is 6.78MHz.
The loop array with a source and a load can be expressed
in an equivalent circuit [2]. The optimum C2, …, Cn, and
RL are determined concurrently using genetic algorithm
(GA) to obtain the maximum efficiency. The C1 is given by
Im(Zin)=0 and RS is set to be Re(Zin), where Zin is input
impedance at the source.
Figure 1. Configuration of 4x4 loop array
Fig. 2 shows the PTE according to a receiver angle
compared with one large loop which has the same size as
the loop array. In the cases of the receiver angle is 0° and
45°, one large loop has the higher efficiencies than the loop
array. However, when the receiver angle becomes 90°, the
one large loop has zero efficiency while the loop array PTE
is still high. The results show that the power is transferred
to an orthogonal receiver by controlling magnetic-field of
loop array.
0 45 90
0
20
40
60
80
100
Effic
ienc
y (%
)
Rotation angle (˚)
Rotated forx-axisy-axisx,y-axisy-axis(One Large Loop)
Figure 2. Power transfer efficiency vs. receiver angle.
Ⅲ. ConclusionThe control of magnetic-field is demonstrated by a 4x4
loop array. By designing the proper values of capacitance
of loop, the magnitude and phase of loop currents can be
controlled and, thus, magnetic-field control is possible.
Therefore, the loop array can transfer power efficiently to
an orthogonal receiver while the conventional orthogonal
loops have zero efficiency. The appropriate capacitances of
loop array are determined using GA for maximum
efficiency.
References
[1] Y. Lim and J. Park, “A Novel Phase-Control-Based
Energy Beamforning Techniques in Nonradiative
Wireless Power Transfer,” IEEE Trans. Power
Electron., vol. 30, no. 11, pp. 6274-6287, Nov. 2015.
[2] B-H Choi, B-C Park, and J-H Lee, “Near-field
Beamforming Loop Array for Selective Wireless
Power Transfer,” IEEE Microw. Wireless Compon.
Lett., vol. 25, no. 11, pp. 748-750, Nov. 2015.
10
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Standing-wave and traveling-wave excitation of a microstrip antenna array
fed by transverse slots on a broad wall of the rectangular waveguide for
linear polarization parallel to the axis o Yuichi Kimura, Fumihiko Nonaka, and Sakuyoshi Saito
Dept. Electrical and Electronic Systems, Graduate School of Science and Engineering, Saitama University
I. Introduction Waveguide slot arrays are commonly used for various
applications such as radars and communication systems. A
longitudinal slot array on a broad wall of the rectangular
waveguide is one of the typical designs for polarization
perpendicular to the waveguide axis. For transverse slot
arrays on a broad wall of the waveguide for polarization
parallel to the axis, reduction of grating lobes becomes a
significant problem because transverse slots are arranged at
a spacing of one guided wavelength that is usually longer
than a wavelength in free space. In order to shorten the slot
spacing, a dielectric-filled waveguide or slow wave
structures in the waveguide are utilized for the transverse
slot arrays [1]. Another solution that uses T-slots arranged
on a ridged waveguide is reported [2].
For that purpose, the authors have proposed a novel
planar array antenna on a broad wall of the rectangular
waveguide for linear polarization parallel to the axis [3]. A
two-element series-fed microstrip antenna (MSA) array
placed on a dielectric substrate, which is excited by a
transverse slot on the broad wall of the waveguide, is used
as an element of the proposed array. It is revealed that a
coupling power ratio of the array element can be controlled
from a few % to around 50% by tuning the dimensions of
the array element [4]. In this paper, array designs of the
proposed array antenna with standing-wave excitation and
traveling-wave excitation are presented.
II. Design of standing-wave excitation array
Figure 1 presents a configuration of the proposed
microstrip antenna array on a broad wall of the rectangular
waveguide with standing-wave excitation. Microstrip patch
antennas on a dielectric substrate are arranged on the broad
wall and two patches are connected by a microstrip line.
The series fed patch array is excited by a transverse slot on
Figure 1. The proposed array with standing-wave
excitation.
(a) E-plane (b) H-plane
Figure 2. Radiation patterns.
the broad wall. In this design, the relative dielectric
constant of the substrate is 2.6. Then, the spacing of the
two patches is approximately 0.62 wavelengths in free
space, which is corresponding to around a half of the
guided wavelength in the waveguide. Thus, the grating
lobes of the transverse slots can be suppressed. Polarization
of the proposed array is parallel to the axis of the
waveguide. In this example, six patches and three
transverse slots are arranged, where inset-feeding method
is used for the patches. One of the ends of the waveguide is
set to the feeding port and the other is terminated by a short.
The coupling power ratio required for the three array
1.2
10.2
22.9
1.2
a
a
a
a
a
a
b
bb
bb
b
lf
lf
lf
33.4
33.4
33.0
Port 1
Short
25.3
126
wls
wls
wls
cd
c d
cd
c d
cd
c d
a = 8.4b = 8.4c = 2.0d = 2.0w = 1.0ls = 10.4lf = 13.2unit:[mm]
θ [deg]
E 2 [d
B]
030
60
90 0
120
150180
30
60
90
120
150
0
-20
-20 -20 0
-20
0
Exp. Co-pol. Sim. Co-pol. Exp. X-pol. Sim. X-pol.
θ [deg]
E 2 [d
B]
030
60
90 0
120
150180
30
60
90
120
150
0
-20
-20 -20 0
-20
0
11
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
elements is 33% for standing-wave excitation array. The
design frequency is 11.2 GHz.
Figure 2 presents the simulated and measured radiation
patterns in E- and H-planes at 11.2 GHz, where the
simulated result is obtained by Ansys HFSS. It is found
that the grating lobes at around 50 degree directions in the
E-plane pattern produced by the transverse slot array
arranged at one guided wavelength are well suppressed.
III. Design of traveling-wave excitation
array Figure 3 presents the proposed microstrip antenna array
on a broad wall of the rectangular waveguide with
traveling-wave excitation. In this design, sixteen patches
and eight transverse slots are arranged. The inset-feeding is
used for the first five elements near the feeding port and
the edge feeding is used for the last three elements. In order
to create uniform excitation distribution, the coupling
power ratio for the element #n numbered from the last is
set to 1/n. Furthermore, the forward beam tilting design is
introduced to suppress the reflection at the feeding port.
The spacing of the transverse slots are slightly apart from
one guided wavelength.
Figure 4 presents the simulated and measured radiation
patterns in E-plane at 11.2 GHz. The grating lobes at
around 50 degree directions produced by the transverse slot
array arranged at one guided wavelength are not observed.
Figure 3. The proposed array with traveling-wave
excitation.
Figure 4. E-plane radiation patterns.
IV. Conclusion A microstrip antenna array fed by transverse slots on a
broad wall of the rectangular waveguide for linear
polarization parallel to the axis is presented. Validity of the
proposed array with standing wave excitation and
traveling-wave excitation is confirmed by simulation and
measurement.
References [1] S. Yamaguchi, et al., “A slotted waveguide array
antenna covered by a dielectric slab with a post-wall
cavity,” IEICE Tech. Rep., vol. 112, no. 7, AP2012-5,
pp. 21-26, Apr. 2012.
[2] S. Mihara and N. Kuga, “T-slot antenna on the ridged
plane of a ridged waveguide,” IEICE Trans. (B), vol.
J95-B, no. 9, pp. 1052-1059, Sep. 2012.
[3] Y. Kimura and F. Nonaka, “A Microstrip Antenna
Array on a Broad Wall of the Rectangular
Waveguide with Polarization Parallel to the Axis,”
Proc. 2013 Korea-Japan Workshop on Antennas and
Propagation, p. 8, Jan. 2013.
[4] F. Nonaka, S. Sakuyoshi Saito, and Y. Kimura,
“Design of a planar array antenna on a broad wall of
the rectangular waveguide for polarization parallel to
the axis with standing-wave excitation,” IEICE Tech.
Rep., vol. 114, no. 354, AP2014-156, pp. 31-36, Dec.
2014. 22.9
1.2
10.21.2
35.0
Port 1
Short25.3
305
35.0
35.0
35.0
16.5
35.0
35.0
35.0
lf1
lf1
lf2
lf3
lf3
lf3
lf3
lf4
ls1
ls1
ls2
ls3
ls4
ls5
ls6
ls7
a
a
a
a
a
a
a
a
a
a
a
a
a
a
a
a
b1
b1b1
b1b2
b2b3
b3b4
b4b2
b2b5
b5b2
b2
c d
c d
c d
c d
c d
c d
c d
c d
c d
c d
w1
w1
w2
w2
w2
w2
w2
w2
2.0
2.0
2.0
2.0
2.0
2.0
2.0
2.0
#1
#2
#3
#4
#5
#6
#7
#8a = 8.4, b1 = 8.4, b2 = 10.8, b3 = 9.0, b4 = 10.4, b5 = 11.2,c = 2.0, d = 2.0, lf1 = 12.0, lf2 = 15.2, lf3 = 13.2, lf4 = 13.6, w1 = 4.4, w2 = 1.0, ls1 = 18.6, ls2 = 9.8, ls3 = 9.4, ls4 = 9.0, ls5 = 8.6, ls6 = 7.9, ls7 = 7.8 unit :[mm]
-90 -60 -30 0 30 60 90-40
-30
-20
-10
0
Angle [deg.]
Rel
ativ
e A
mpl
itude
[dB
]
Exp. Co-pol. Sim. Co-pol. Exp. X-pol. Sim. X-pol.
12
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
A Frequency-Reconfigurable Dipole Antenna Using a Tapered Impedance
Matching Structure
Jang-soon Park, Jun-Bong Ko, and o Dongho Kim
Department of Electronic Engineering, Sejong University
I. IntroductionRapid appearance of diverse wireless communication
services has continuously been increasing the demand for
frequency reconfigurable antennas. For wideband operation,
good impedance matching between an input port and an
antenna’s radiating part is necessary. Accordingly, we
propose a versatile wideband balun structure enables good
impedance matching in a broad frequency range.
II. Design and experimentThe geometry of a proposed antenna is given in Fig. 1.
The antenna consists of a microstrip line, a parallel plate
line, a tapered balun, and two radiating arms with inserted
varactor diodes, which is fed by a 50 • coaxial connector.
The lines and arms have been etched on each side of a 1.52
mm thick commercial Taconic RF-35 dielectric substrate.
In order for frequency scan in a wide frequency range,
good impedance between the input port and the dipole
antenna is necessary. To do that, the tapered parallel plate
line has been introduced as show in Fig. 1, which not only
transfers waves to the dipole antennas (arms) with low
reflection but changes field distribution suitable for dipole
radiation. In fact, the input impedance varies from 50 • to
196 • .
The varactor diodes (SMV-1405) from Skyworks have
been used to electrically scan the resonant frequency of the
antenna. A 5 pF DC blocking capacitor and a 20 nH RF
choke inductor have been used to prevent the undesirable
influence of DC and RF signals, respectively.
x y
z
Varactor diode
RF choke
DC blolck
Parallel plate line
Figure 1. Fabricated antenna
2.5 3.0 3.5 4.0 4.5-35
-30
-25
-20
-15
-10
-5
0
S11
[dB
]Frequency [GHz]
0v Mea. 30v Mea. 0v Sim. 30v Sim.
Figure 2. Comparison of the simulated and measured
reflection coefficient with different values of bias voltage.
The simulated and measured reflection coefficient is
shown in Fig. 2, which proves our antenna successfully
hops from 3.35 GHz to 3.78 GHz when the bias voltage
switches from 0 V to 30 V.
III. ConclusionWe have proposed the tapered balun structure which
shows good impedance matching in a wide frequency
range and can be used in various antenna applications.
AcknowledgementThis work was supported by Institute for Information &
Communications Technology Promotion (IITP) grant
funded by the Korea government (MSIP). [R-20150224-
000291, Development on Semi-conductor based Smart
Antenna for Future Mobile Communications]
References
[1] P. Y. Qin, A. R. Weily, Y. J. Guo, T. S. Bird, and C.
H. Liang, “Frequency reconfigurable quasi-Yagi
folded dipole antenna,” IEEE Trans. Antennas
Propag., vol. 58, no. 8, pp. 2742-2747, Aug. 2010.
13
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
A Dual-Band GPS Antenna Integrated Inside a Military Helmet
Nu Pham and oJae-Young Chung
Seoul National University of Science and Technology
Ⅰ. IntroductionRecently, the capability of precisely locating and
tracking a device, vehicle or human has received more and
more attention. However, the positioning Error of a
conventional Global Navigation Satellite system (GNSS) is
limited to several meters due to multipath, ionospheric and
trospheric delays. One method to mitigate such errors is
using a dual-band operation system implemented with a
dual-band, dual-circularly polarized (CP) antenna.
Here we present a design of dual-band antenna
integrated in a military helmet for tracking a soldier in the
field. The standalone antenna was presented in AWAP2015,
Bangkok, Thailand [1]. The contents of this paper focus on
the implementation of the antenna inside an helmet in a
limited installation area and with possible head effects.
Ⅱ. Antenna Installation and SimulationThe standalone dual-band antenna was designed to
operate at L1 (1.57 GHz) and L2 (1.23 GHz) bands of
Global Positioning System (GPS). Figure 1 depicts the
antenna integrated in the military helmet. This microstrip
GPS antenna has a single feed fed by the side of the PCB
to ease the connection between the antenna and an external
receiver. Also, it is compact and low-profile suitable for
installation. The antenna is as small as 73mm×73mm×6.4
mm, corresponding to 0.29• ×0.29• ×0.026• at 1.227 GHz.
In the full-wave simulation model, a 3D model of
helmet was imported from a CAD file. In addition, a
spherical head phantom was designed to investigate the
head effect on the antenna performance. The head phantom
consists of three materials: skin, skull and brain tissue, and
their material properties are assigned based on [2].
Figure 2 shows the comparisons of antenna reflection
coefficient (S11) and axial ratio (AR) when the antenna is
in the free-space, with helmet, and with helmet and head
phantom. It can be seen that the S11 and AR are shifted to
the higher frequency as the helmet exists. On the other
hand, the effect of phantom is trivial due to the large
ground isolating the high loss head phantom.
Figure 1. GPS antenna mounted in helmet simulation.
Figure 2. Comparisons S11 and axial ratio.
Ⅲ. ConclusionThe proposed method of attaching antenna inside the top
of helmet takes an important role on remaining operation
of standalone antenna, adapt with wearable requirement in
military equipment.
Acknowledgement
This work was supported by the Basic Science Research
Program through the NRF Korea funded by the Ministry of
Science, ICT & Future Planning (No.
2013R1A1A1005735).
References
[1] N. Pham and J.-Y. Chung, "A miniaturized circular
patch antenna for dual-band GPS application,"
AWAP2015, Bangkok, Thailand, 2015.
[2] http://niremf.ifac.cnr.it/tissprop/htmlclie/htmlclie.php
1.1 1.2 1.3 1.4 1.5 1.6 1.7-50
-40
-30
-20
-10
0
Frequency [GHz]
S11
(dB
)
StandaloneHelmetHelmet+Phantom
1.1 1.2 1.3 1.4 1.5 1.6 1.70
3
6
9
12
Frequency [GHz]
AR
(dB
)
StandaloneHelmetHelmet+Phantom
Helmet
Patch Antenna Large ground
BrainSkin
Skull
14
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Helmet Folded Dipole Antennas
Naobumi Michishita, Naoto Nishiyama, and Hisashi Morishita
National Defense Academy
I. IntroductionHelmet antennas have been developed for integrating
into the helmet for disaster prevention [1]. To reduce the
degradation of antenna characteristics due to the human
head, an inverted-F antenna with a copper ring structure
has been proposed [2]. However, the inverted-F antenna
has the narrow bandwidth. This paper presents the folded
dipole antenna with the slit-loaded copper ring structure.
II. Antenna CharacteristicsFigure 1 shows the simulation models of the proposed
folded dipole antenna with human head. The antenna
element is arranged at 5 mm from the edge of the
hemispherical dielectric shell with a diameter of 125 mm.
To achieve the impedance matching, the widths of the feed
and non-fed arms of the dipole are 3 mm and 12 mm,
respectively. Fig. 1(b) shows the slit-loaded copper ring
structure.
Figure 2 shows the simulated VSWR characteristics. The
relative bandwidth at VSWR = 3 becomes 2.7% and 1.9%
with and without the slit, respectively. Figure 3 shows the
radiation efficiencies. The efficiency of 8.6% can be
improved by loading the slit. Figure 4 shows the simulated
10 g average local SAR distributions. The unwanted
radiation toward the human head can be suppressed and
SAR value is reduced.
III. ConclusionThe proposed helmet folded dipole antenna with slit-
loaded copper ring structure has high radiation efficiency
and low SAR value.
References
[1] T. Nakao, H.T. Nguyen, M. Nagatoshi, and H.
Morishita,“Fundamental study on curved folded
dipole antenna,” IEEE AP-S Int. Symp., Chicago, IL,
pp.1-2, July 2012.
[2] N. Nishiyama, N. Michishita, and H. Morishita,
“Low-frequency inverted-F antenna on annular
ground plane,” IMWS-Bio, Taipei, Taiwan, pp.143-
144, Sept. 2015.
z
yx
111
22
Human head
[Unit: mm]
15
Dielectric shell
Slit
Copper
(a) (b)
Figure. 1 (a) Helmet folded dipole antenna with human
head. (b) Slit-loaded copper ring structure.
140 145 150 155 1601
2
3
4
56
7
8
9
10
w/o Slitw/ Slit
V
SWR
Frequency [MHz]
Figure 2 VSWR characteristics.
140 145 150 155 1600.0
0.2
0.4
0.6
0.8
1.0
w/o Slitw/ Slit
Rad
iati
on e
ffic
ienc
y
Frequency [MHz] Figure 3 Radiation efficiencies.
1[W/kg]
0
(a) (b)
Figure 4 10 g average local SAR distributions of (a) with
and (b) without slit.
15
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Design of Frequency Selective Spiral Slot located in Near-Field
of Wireless Power Transfer System by Eigenmode Analysiso Kunio Sakakibara, Kei Firdaus, Nobuyoshi Kikuma
Nagoya Institute of Technology
Ⅰ. IntroductionA transmitting circuit of a wireless power transfer
system is shielded by a metal case to prevent noise
emission. On the other hand, a receiving circuit equipped
in cars or mobile devices are located outside of the shield
case. Therefore, the transmitting and receiving antennas are
separated by the metal plate. A frequency selective spiral
slot is proposed to be cut on the metal plate. Only the
transmitting power in the operating frequency of the spiral
slot can transmit to the receiving antenna. The spiral slot is
designed by eigenmode analysis individually from the
wireless power transfer system. The simulated and
measured performances are demonstrated in this paper.
Ⅱ. AnalysisTransmitting and receiving helical antennas are
separated by an infinite ground plane with the frequency
selective spiral slot as shown in Fig. 1. The two areas of the
transmitting and receiving antennas are electrically
connected only through the spiral slot. The operating
frequency is 50MHz band. To reduce the physical size of
this system, only one spiral slot is used, although popular
frequency selective surfaces (FSS) are composed of
periodic structure. The diameters of the helical and the
spiral slot are 120mm and 112mm, respectively. The
ground plane with the spiral slot is at the center between
the helical antennas separated by 40mm, where the stored
electromagnetic field of the wireless power transfer system
distributes. The resonant frequency of the spiral slot is
designed to be the same as the wireless power transfer
system by eigenmode analysis of finite element method.
The simulated resonant frequency was 59.75MHz.
To demonstrate the performance of the proposed system,
the analysis model in Fig. 1 was fabricated for
measurements. The photograph of the spiral slot on the
copper ground plane is shown in Fig. 2. The size of the
ground plane was 300mm square. The simulated and
measured transmission property is shown in Fig. 3. The
simulated resonant frequency was 59.5MHz which is
almost the same with the eigenmode analysis. The
measured transmission was −4.7dB (34%).
Ⅲ. ConclusionThe frequency selective spiral slot is designed by
eigenmode analysis independent on the wireless power
transfer system, therefore, the resonant frequencies of the
wireless power transfer systems and the spiral slot may be
perturbed. However, the transmission property was still
conserved even in the near field of the wireless power
transfer system.
Figure 1. Analysis model.
Figure 2. Frequency selective spiral slot. (25 windings)
Figure 3. Transmitting property.
16
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
PIFA antenna for UWB applicationso Do-Gu Kang and Jaehoon Choi
Department of Electronics and Computer Engineering, Hanyang University
[email protected] (corresponding author)
Ⅰ. IntroductionRecently, ultrawideband (UWB) communication
systems have received much attention because of supporting high data rate and low power consumption [1].Due to the limited area available for an antenna, UWB antenna should have compact size and low height [2]. Although the planar inverted-F antenna (PIFA) has compact size and low height, the PIFA is not suitable for UWB antenna due to its narrow bandwidth.
To overcome this problem, PIFA antenna with slotted ground plane for UWB applications is proposed. The proposed antenna has stable and near omnidirectional radiation characteristic.
Ⅱ. Antenna design and simulation resultFigure 1 shows the geometry of the proposed antenna.
The proposed antenna consists of a PIFA and ground plane with a slot. The PIFA is placed on the top of an FR4 substrate (•r = 4.4) with 1 mm thickness. The wideband impedance matching of the PIFA is realized by adding aslot on the ground plane. The ground plane is located on the bottom of the substrate and has a total size of 30 mm ×50 mm.
As shown in Figure 2, a -10 dB reflection coefficient bandwidth of the proposed antenna satisfies the full UWBfrequency range (3.1 GHz - 10.6 GHz).
Figure 3 illustrates the simulated radiation patterns. The proposed antenna has stable and near omnidirectional radiation patterns. Peak gains are 4.53 dBi, 3.55 dBi, 5.14 dBi, 6.08 dBi, 5.16 dBi at 3.4 GHz, 4.19 GHz, 4.88 GHz, 7.39 GHz, 10.2 GHz, respectively.
Ⅲ. ConclusionIn this paper, a PIFA antenna for UWB applications is
proposed. The proposed antenna has the wide -10 dB reflection coefficient bandwidth (3.08 GHz - 10.67 GHz)
Figure 1. Geometry of the proposed antenna.
Figure 2. Simulated reflection coefficient.
(a) (b)Figure 3. Simulated radiation patterns (a) xy-plane, (b) yz-plane.
satisfying the UWB frequency range by adding a slot on the ground plane. The antenna provides stable and near omnidirectional radiation for UWB applications.
References[1] G.-P. Gao, B. Hu, and J.-S. Zhang, “Design of a
miniaturization printed circular-slot UWB antenna by the half-cutting method,” IEEE Antennas and Wireless Propagation Letter, vol. 12, pp. 567-570,2013.
[2] A. Foudazi, H. R. Hassani, and S. M. A. Nezhad, “Small UWB planar monopole antenna with addedGPS/GSM/WLAN bands,” IEEE Transactions on Antennas and Propagation, vol. 60, no. 6, pp. 2987-2992, 2012.
AcknowledgementThis work (Grants No. C0331675) was supported by
Business for Cooperative R&D between Industry, Academy, and Research Institute funded Korea Small and Medium Business Administration in 2015.
17
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Wideband Sequential-Rotation Arrays with Circularly Polarized Patch
Radiators using Anisotropic Metasurface Sarawuth Chaimool1, Prayoot Akkaraekthalin2, and Kwok L. Chung3
1Udon Thani Rajabhat University, Thailand 2King Mongkut’s University of Technology North Bangkok, Thailand
3Qingdao Technological University, China
I. Introduction Recently metasurface used for microwave antenna
applications becomes popular after supporting theories [1].
Novel types of metasurface for various applications with
significant improvements on microwave antennas were
reported [2-8]. The high-gain CP antenna was successfully
produced by using a grounded anisotropic metasurface in [2].
Simultaneous enhancement of gain and bandwidths of LP
and CP patch antennas were demonstrated in [3-5].
Polarization conversion using novel metasurfaces was
presented in [6-7]. Metasurfaces used for improving front-to-
back radiation ratio of slot/aperture antennas were illustrated
in [8-9]. More recently, performance enhancement and
sidelobes suppression of a conventional low-cost CP patch
array using a thin dielectric layer with metallic patterned cells
designated as the anisotropic metasurface (AMTS) have been
proposed in [10]. In this paper, we further study the sidelobe
levels suppression of the CP array by using various types of
thin AMTS.
II. Performance Enhancement of CP Array
A. Geometry of the original array and AMTS Figure 1 shows the geometry of the low-cost CP patch
array with the addition of the AMTS [10], where a layered structure was used for the wideband feed-network, 2-by-2 CP patches and the thin (0.007λo) metasurface. The metasurfaced CP array has an overall height of only about 0.073 λo.
B. Sidelobes Suppression In addition to the simultaneous enhancement of boresight
gain, gain bandwidth, axial-ratio and impedance bandwidths, suppression of sidelobe levels from both the co-polar (LHCP) and x-polar (RHCP) patterns were also achieved as evident in Figs. 2 and 3. The CP patches have an element spacing of 0.8 λo, the maximum sidelobe level (SLL) was recorded as high as -9.1 dB (average of max SLLs at two principal planes) at 2.45 GHz, as shown in Fig. 2. After mounting the AMST, the radiation patterns shown in Fig. 3 demonstrate an average reduction of SLLs by 4.4 dB. This is one of the reasons that explain the boresight gain enhancement of about an octave.
Figure 1. Geometry of CP patch array with AMTS.
Figure 2. Radiation patterns of CP array without AMTS.
Figure 3. Radiation patterns of CP array with AMTS-1.
III. Further Sidelobe Levels Suppression In this study, we aim to further suppress the sidelobe
levels of the CP array by simply reconfiguring the metasurface, rather than a complicate feed-network and or the element spacing. Two types of anisotropic metasurface have been investigated. Their performances are compared
-30dB
-20dB-10dB
0dB
90-90
180
= 0
xz-plane 2.45 GHz
LHCP RHCP
z
x
HPBW = 31.2Max SLL = -9.39dB at -62
yz-plane 2.45 GHz
LHCP RHCP
z
y -30dB
-20dB
-10dB0dB
90-90
180
= 0
HPBW = 30.8Max SLL = -8.76dB at 58
-30dB
-20dB-10dB
0dB
90-90
=180
0
xz-plane 2.45 GHz
LHCP RHCP
z
x
simulation measurement LHCPRHCP
HPBW = 28.6Max SLL = -13.4dBat -56
yz-plane 2.45 GHz
LHCP RHCP
z
y
simulation measurement LHCPRHCP
-30dB
-20dB
-10dB0dB
90-90
=180
0
HPBW = 29Max SLL = -13.6dBat -56
18
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
with AMTS-1, viz., the original one presented in [10]. Fig. 4 shows the geometries whereas Fig. 5 presents their SLL suppression performances. As can be seen, the inner 4 cells of each AMTS were modified. In AMTS-2, we replaced the diagonal strips by the 10-mm square rings whereas AMTS-3 has detached diagonal strips.
(a) (b)
Figure 4. Geometries of anisotropic metasurfaces: (a) AMTS-2, (b) AMTS-3.
(a) xz-plane
(b) D-plane
Figure 5. SLL performances of CP array after mounting different AMTS.
IV. Concluding Remarks New types of anisotropic metasurface for performance
enhancement of CP patch array are proposed in this paper. Further suppression of sidelobe levels without sacrificing the wideband and high-gain performance is achieved. The new metasurfaces can suppress up to 3 dB more in the principal planes but up to 10 dB in the diagonal plane. More results will be presented and discussed in the conference.
References [1] C. L. Holloway, E.F.Kuester, J.A.Gordon, J.O’Hara, J.Booth,
and D.R.Smith, “An overview of the theory and applications of metasurfaces: The two-dimensional equivalents of metamaterials,” IEEE Antennas Propag. Mag., vol.54, (2), pp.10–35, Apr.2012.
[2] G. Minatti, S. Maci, P. De Vita, A. Freni and M. Sabbadini, “A circularly-polarized isoflux antenna based on anisotropic metasurface,” IEEE Trans. Antennas Propag., vol.60, (11), pp.4998-5009, 2012.
[3] K. L. Chung and S. Chaimool, “Diamagnetic metasurfaces for performance enhancement of microstrip patch antenna,” 5th European Conference on Antenna and Propagation, pp. 55-60, EuCAP 2011, Apr 11-15, 2011, Rome, Italy.
[4] S. Chaimool, K. L. Chung, and Prayoot Akkaraekthalin, “Simultaneous gain and bandwidths enhancement of a single-feed circularly polarized patch antenna using a metamaterial reflective surface,” Progress In Electromagnetics Research B, Vol. 22, pp. 23-37, 2010.
[5] M. H. Ullah and M. T. Islam, “A new metasurface reflective structure for simultaneous enhancement of antenna bandwidth and gain,” Smart materials and Structures, Vol. 23, (8), 085015, 2014.
[6] H. L. Zhu, K. L. Chung, X. L. Sun, S. W. Cheung and T. I. Yuk, “CP Metasurfaced Antennas Excited by LP Sources,” IEEE Antennas and Propagat., Society Intern Symp 2012.
[7] H. L. Zhu, S. W. Cheung, K. L. Chung, and T. I. Yuk, “Linear-to-circular polarization conversion using metasurface,” IEEE Trans Antennas and Propagat., Vol. 62, (9), pp. 4615-4623, 2013.
[8] S. Sarawuth, C. Rakluea and P. Akkaraekthalin, “Mu-near-zero metasurface for microstrip-fed slot antennas,” Appl Phys A, vol. 112(3), pp.669-675, 2013.
[9] K. L. Chung and S. Kharkovsky, “Metasurface-loaded circularly-polarized slot antenna with high front-to-back ratio,” Electronics Letters, Vol. 49, no. 16, pp. 979-981, Aug. 2013.
[10] K. L. Chung, S. Chaimool and C. Zhang, “Wideband subwavelength-profile circularly-polarized array antenna using anisotropic metasurface,” Electronics Letters, Vol. 51, (18), pp. 1403-1405, Sep. 2015.
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The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Electrically small spherical antennas using 3D printing technology Myeongjun Kong, Geonyeong Shin, and Ick-Jae Yoon
Dept. of Electrical Engineering, Chungnam National University, Daejeon, Korea
Ⅰ. Introduction For the past decades, there has been much interest on the
theoretical analysis of the radiation properties of electrically
small antennas and practical realization of them. The
theoretical bound is represented by the radiation quality
factor Q versus the electrical size of the antenna ka, where k
is the free space propagation constant and a is the radius of
the imaginary sphere enclosing the antenna. Among the
many practical antenna designs, a folded spherical helix
(FSH) dipole antenna is one of the well-known designs
approaching the low Q bound [1]. However, such the design
has a limit in miniaturization since the wires are physically
overlapped each other as the ka becomes smaller. Under this
background, this paper designs an FSH dipole antenna made
of thin and wide metal strips instead of thick wires, which
enables much smaller designs approaching the Q bound.
Ⅱ. FSH dipole antenna made of thin and
wide metal strip At high frequency, a thick wire can be electrically
replaced by a thin and wide metal strip with the same
circumference. It becomes practically available to design an
antenna with very small ka then when such the metal strip is
utilized for the excitation and folding arms in an FSH
structure.
To verify the proposed concept, we first design an FSH
antenna with ka=0.38 using a thick wire [1] and a thin and
wide metal strip with the same circumference, respectively.
It is found that the two antennas show about the same
radiation properties such as the Q value, input impedance
and radiation efficiency. Consequently, we design an FSH
antenna with ka=0.21 using the thin and wide metal strip. It
is worth to note that that the multiple folding arms should be
overlapped when the wires with the same circumference are
used. One may use thinner wire to avoid the overlapping,
but it will result in low radiation efficiency. The increased
number of arms with thinner wire for stepped-up radiation
efficiency will also cause physical overlapping. Thus, even
smaller FSH antenna can be designed without the physical
limitation using the proposed method of utilizing metal strip.
In Fig. 1, it is observed from the full wave EM simulation
that the designed antenna with ka=0.21 resonates at 301
MHz with radiation efficiency of 89.7%. It is also found that
it approaches the low Q bound. The proposed antenna is
fabricated using the 3-D printing technology and copper
coating process. The radiation characteristics together with
the measurement results will be reported at the conference.
(a)
(b)
Fig. 1. Simulated results of the designed FSH antenna
made of thin and wide metal strip (ka=0.21). (a) Reflection
coefficient. (b) Radiation efficiency.
Ⅲ. Conclusion This paper shows that even smaller FSH dipole antenna
approaching the low Q bound can be designed using the
proposed thin and wide metal strip structure which replaces
the thick wires in the original design [1]. The proposed
design can be fabricated using the commercialized 3-D
printing technology and copper coating process.
References [1] S. R. Best, “The radiation properties of electrically
small folded spherical helix antennas,” IEEE Trans.
Antennas Propag., vol. 52, no. 4, pp. 953-960, April,
2004.
20
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Beamwidth Reconfigurable Array Antenna
Without Power Loss Using the Switched Couplero Soon-Soo OH, o Dong-Woo Kim, o Tae-Hyung Kim, and * Chi-Hyung Ahn
o Chosun University, * Korea Railroad Research Institute
Ⅰ. IntroductionThe RFID was proposed to be a good candidate for train
position detection where the beamwidth was controlled for
the optimum detection depending on the speed of train [1].
In this paper, the beamwidth reconfigurable array antenna
was proposed. Compare with the previous technique [1],
the beamwidth was controlled in three-way switched
coupler.
Ⅱ. Coupler and Antenna DesignThe proposed coupler is shown in Fig. 1. The input
port is Port 1 and the output port is Port2, Port3, and
Port 4. The Port 5 and Port 6 is isolated port. The
central frequency is 915 MHz. The dielectric
constant and height of the substrate is 4.5 and 1.6mm.
Fig. 1. Proposed coupler using the three-way output.
The four switches connected to the ground is place
on the vertical branch of the coupler as shown in Fig.
1. When they all are open, the transmission
coefficient was S21 = -4.69 dB, S31 = -5.28 dB, S41 = -
4.69 dB as shown in Fig.2. Meanwhile, when two
switches are open, the transmission coefficients are
S21 = -20.2 dB, S31 = -3.11 dB, S41 = -3.77 dB. When
all switch are closed, the transmission coefficients
are S21 = -17.4 dB, S31 = -0.34 dB, S41 = -17.4 dB.
Therefore, depending on the switch status, the power
could be delivered to three ports, two ports or only
one port. Compared with the previous technique of
cutting the branch, the proposed technique has the
almost zero loss.
Fig. 2. Reflection and transmission coefficients for
three-way coupler.
This coupler has been connected to the antenna
array with three radiating elements. The beamwidth
was controlled by switching the connection to the
ground, which will be presented in the conference.
Ⅲ. ConclusionIn this paper, the beamwidth controlled array antenna
was proposed with the three-way reconfigurable coupler.
The proposed array antenna could be used for the
application of the train position detection.
Acknowledgement
This research was supported by a grant from the Advanced
Technology Center R&D Program funded by the Ministry
of Trade, Industry & Energy of Korea (10048475).
References
[1] C.-H. Ahn, B.-K. Cho, S.-H. Ryu, S.-S. Oh, “Design
of beam-forming reader antenna for train position
detection using RFID,” Journal of the Korean
Society for Railway, vol. 18, no. 2, pp. 105-110, Apr.
2015.
21
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Ⅰ. Introduction With the development of wireless communications,
antennas capable of tunable operation have been required
for various applications. A ferrite-loaded antenna is a good
candidate for this requirement. It is well-known that the
electromagnetic properties of the ferrites can be varied by
applying a static magnetic field. This feature can be
utilized to develop multifunctional antennas. In this talk,
mode splitting behavior of biased ferrite resonator antennas
will first be addressed [1]. Then, three tuning performances
(frequency, bandwidth, and polarization) of the antenna
will be discussed.
Ⅱ. Mode Splitting Behavior of Ferrite-
Loaded Resonator Antennas Theoretical models of cylindrical resonator
antennas with a static magnetic field along the z-axis are shown in Fig. 1.
It is known that for the analysis of cylindrical resonators, it
is possible to introduce a simplification which leads to
conventional eigenvalue equations [2]. Mode splitting
behavior of the cylindrical resonator antennas is verified by
mode matching technique. This technique will be
addressed. In addition, mode splitting behavior (see Figs. 2
and 3) enables three tuning performances: frequency,
bandwidth, and polarization. These features will be also
discussed.
III. Conclusion Theoretical models of ferrite-loaded resonator antennas
are found to be efficient to explain the HE11δ mode splitting
behavior. In addition, this unique feature provides various
functions: frequency and bandwidth tuning, and
polarization switching performances. This indicates that
the proposed antennas are suitable for multifunctional
antennas.
References [1] B.Y. Park, T.W. Kim and S.O. Park, “Analysis of
HE11δ mode splitting behavior of cylindrical ferrite
resonator antenna,” Korean Institute of Electro-
magnetic Engineering and Science Summer Sympo-
sium, Jeju, Korea, 2015.
[2] D. Kajfez and P. Guillon, Dielectric Resonators,
NOBLE, Tucker, Georgia, USA, 2007
Byeong-Yong Park, Tae-Wan Kim, and Seong-Ook Park*, Senior Member, IEEE School of Electrical Engineering, Korea Advanced Institute of Science and Technology, Daejeon, Korea
(a) (b)
Figure. 1. Theoretical models (a) ferrite resonator
antenna, (b) hybrid resonator antenna.
Figure. 2. Frequency response of cylindrical ferrite
resonator antenna for various DC magnetic bias.
Figure. 3. Frequency response of cylindrical hybrid
resonator antenna for various DC magnetic bias.
Analysis of Mode Splitting Behavior for Cylindrical Ferrite Resonator
Antenna
22
- 23 -
AWAP 2016Asian Workshop on Antennas and Propagation
Plenary Talk and Invited Session
28th January, 2016Thursday
Plenary Speaker Biography
Toshikazu Hori received the B.E., M.E. and Dr. Eng. degrees in electrical engineering from Kanazawa University, Japan, in 1974, 1976 and 1993, respectively. In 1976, he joined the Electrical Communications Laboratories, Nippon Telegraph and Telephone Public Corporation (now, Nippon Telegraph and Telephone Corporation, NTT Corp.). Since then, he has been engaged in the research and development of antennas for satellite, cellular and microcellular mobile, and broadband wireless communication systems. In 2001, he moved to the University of Fukui, and is currently a Professor at the Graduate School of Engineering. His current research interests lie in the area of antennas and propagation for wireless broadband systems, especially, broadband antennas and meta-surfaces. Prof. Hori served as the Honorary Chair of AWAP2014 in Kanazawa, Japan. He also served as the Vice-Chair and the Chair of the IEEE AP-S Japan Chapter from 1999 to 2000, the Vice-Chair and the Chair of the IEEE AP-S Nagoya Chapter from 2007 to 2010, the Vice-Chair of IEEE Nagoya Section from 2013 to 2014, respectively. He was also the Guest Editor-in-Chief of four special issues in IEICE Trans., the Vice-Chair of ISAP2004 in Sendai, the TPC Chair of ISAP2007 in Niigata, the Vice-Chair and the Chair of the Technical Committee on Antennas and Propagation (TC-AP) of IEICE Japan from 2005 to 2009, and the Representative of IEICE Hokuriku Section, respectively. He is now the Advisory Committee Member of the TC-AP of IEICE Japan. He is a Fellow of the IEEE, and is also a Fellow of the IEICE Japan.
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Low-Profile Design of Meta-Surface
with Frequency Selective Surface and Its Application Toshikazu Hori
University of Fukui, Japan
[email protected], [email protected]
Abstract An artificial magnetic conductor (AMC) and a high
impedance surface (HIS) have the perfect magnetic
conductor (PMC) characteristics at the specific frequency.
Meta-surfaces with an arbitrary reflection phase have been
studied extensively. And, these meta-surface technologies
have also applied to the antennas and propagation area.
The meta-surface composed of both frequency selective
surface (FSS) and the ground plane was proposed for an
easy configuration. By utilizing the spatial filter
characteristics of FSS well, a meta-surface with desired
reflection characteristics can be realized simply.
In this plenary talk, a low-profile design method using
both FSS and the ground plane is presented, and its
application to the antennas and propagation area are
introduced. When the spatial filter characteristics of FSS is
given, the relational equation between the reflection phase
θs and the thickness h of the meta-surface is derived by
using approximate optical ray theory. For an arbitrary
frequency f and thickness h, the meta-surface with the
desired reflection phase can be designed using this
relational equation. Here, the design method of a meta-
surface is made clear by using this relational equation, and
several design examples are introduced.
(1) Low-profile design of AMC
A loop slot type FSS which is a band-pass filter, a loop-
type FSS which is a band-rejection filter and a patch type
FSS with low-pass filter characteristics in the similar size
are considered as the FSS for an AMC with PMC
characteristics at the specific frequency. The thickness h
between FSS and the ground plane with PMC
characteristics at the arbitrary frequency f is derived from
the above-mentioned relational equation. As a result, the
low-profile design method with wider PMC relative
bandwidth is made clear. And, it is shown that the meta-
surface using loop-type FSS (as shown in Fig.1) is
excellent in PMC relative bandwidth, a low profile, and the
directive gain when employing as the reflector of a dipole
antenna.
(2) Design of meta-surface with the PMC characteristics
Here, a meta-surface with the frequency independent
PMC characteristics is considered while an AMC has PMC
characteristics at the specific frequency. In order to realize
a meta-surface with the frequency independent PMC
characteristics, the spatial filter characteristics of FSS is
derived from the above-mentioned relational equation. And,
the achieved possibility is verified.
(3) Design of meta-surface for polarization conversion
The principle of the meta-surface for polarization
conversion from linearly polarized wave to circularly
polarized wave is shown. The design method of the meta-
surface for polarization conversion using the patch type
FSS is also introduced. As a result, it is shown that the
meta-surface for polarization conversion has broadband
axial ratio characteristics. Next, the ideal filter
characteristics of FSS is derived from the above-mentioned
relational equation. And, the configuration of FSS suitable
to the meta-surface for polarization conversion is also
made clear.
Figure 1. Configuration of meta-surface with loop-type
FSS.
(Plenary Talk)
25
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Surface moisture content sensor detecting mutual coupling magnitude
between parallel and perpendicular meander line dipole antennas
Chattapon RIENTHONG1, Chainarong KITTIYANPUNYA 2, and Monai KRAIRIKSH1**
1Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang, 1 Chalongkrung Rd., Ladkrabang,
Bangkok10520, Thailand
E-mail: [email protected], [email protected]
Abstract This research presents a surface moisture content sensor detecting mutual coupling magnitude between parallel and perpendicular
meander line dipole antennas. It operates at 245 MHz by detecting mutual coupling magnitude when dielectric properties of soil changes
corresponding to moisture content. The result can be used for creating a graph that can predict a moisture content of soil by using difference
of mutual coupling of parallel and perpendicular meander line dipole antennas. From this technique, an accurate result can be obtained and it
is essential to the effective irrigation system.
Keyword: Soil moisture content sensor, mutual coupling, parallel and perpendicular coupling, meander line dipole antennas.
1. Introduction
Di e l ec t r i c p ro p e r t i e s i s an i n d i c a to r t o
ch a r ac t e r i z e a ma t e r i a l . So i l mo i s tu re co n t en t i s
i mp o r t an t t o t h e y i e ld o f p l an t a t i o n an d i t i s
n ec e s sa r y t o kn o w a c cu r a t e l y . M an y r e se a rch h a v e
b een co n d u c t ed to d e v e l o p so i l mo i s tu re co n t en t
sen so r , e . g . [ 1 ] wh ich ad o p ted u l t r a - wid eb an d r ad ar .
Th e ad van ta g e an d d i sad v an t a ge o f ea ch t e ch n iq u e
i s c r i t i c a l l y r e v i e wed in [ 2 ] . Fo r a wid e a r e a , a
sen so r t h a t c an me a su re i n r ea l t i me i s d es i r ab l e . I t
mu s t p ro v id e f a s t r e su l t s an d d o n o t n e ed to p lu n g e
in to t h e so i l . Wh i l e u s in g o n l y o n e an t en n a i s
e f f e c t i ve [ 3 ] , i t n e ed s a c o mp l e x me a su r e men t . Th e
co s t e f f e c t i v e f r e e sp a c e t e ch n iq u e , i . e . u s in g
co u p l ed an t en n a s an d r e f l e c t i o n co e f f i c i en t [ 4 ]
wh i ch r eq u i r e s a d i r e c t i o n a l co u p l e r wa s u s ed fo r
d e t e r min in g d i e l ec t r i c p r o p er t i e s a t t h e su r f a c e o f a
ma t e r i a l . Th e wo r k in [ 5 ] wa s d e v e lo p ed fo r i n - s i t umo n i to r in g mo i s tu r e co n t en t o f p ad d y wi th o u t u s in g
a d i r e c t i o n a l co u p l e r b y me a su r in g co u p l ed s i gn a l s
f ro m p a r a l l e l an d p e rp e n d icu l a r d ip o l e an t en n a s .
Th e d i e l e c t r i c p ro p er t i e s o f p ad d y wa s d e t e r min e d
f ro m in t e r s ec t io n o f mu tu a l i mp ed an c e f ro m p a ra l l e l
an d p e rp en d i cu l a r d ip o l e a n t en n as o n a su r f a c e cu r v e .
Fo r t h e d i e l e c t r i c p ro p e r t i e s d e t e r min a t io n a t t h e
su r f a c e wi th o u t u s in g a d i r ec t i o n a l co u p l e r , t h e
co u p l ed s i gn a l s f ro m p ar a l l e l an d p e rp en d icu l a r
d ip o l e an t en n as c an b e ap p l i ed . Th i s can b e
a cco mp l i sh ed b y mo d i fy i n g e xp r es s io n s fo r mu tu a l
i mp ed an c e b e t we en th e d ip o le an t en n a s i n [ 5 ] . B y
p ro p e r l y mo d i fy in g th e e xp r es s io n s , t h e su r f a c e
cu r v e fo r d e t e r min a t io n o f 𝜀𝜀𝜀𝜀𝑟𝑟𝑟𝑟′ an d 𝜀𝜀𝜀𝜀𝑟𝑟𝑟𝑟" f ro m me asu r ed
co u p l ed s i gn a l f ro m p a ra l l e l an d p e rp en d i cu l a r
d ip o l e an t en n a s can b e u t i l i zed .
2. Principle of operation The work in [5] was proposed for plunging the sensor
in the material of interest. However, for convenience in
measuring large amount of data in wide area, this work
placed the sensor on the surface of the material of interest
(Invited paper)
26
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
which is soil moisture content. By using the data of soil
from [6], and modifying the calculation for half space (air
and soil) from full space (soil) of the mutual coupling from
[5], the mutual coupling can be calculated. Furthermore, it
was found that low mutual coupling was obtained at
microwave frequency. Hence, lower frequency like 245
MHz (citizen band) was used. In this regard, meander line
dipole antenna [7] was used instead of straight dipole
antenna to miniature the size of the sensor. Note that one
was for parallel polarization whereas the other one was for
perpendicular polarization. The width of the sensor is 30
cm.
3. Results An oscillator from a transmitter of 245 MHz was used
for transmitting signal via a transmitting antenna. The
receiving antennas consisting of parallel and perpendicular
polarized meander line dipole antennas which an RF switch
was used for selecting the polarization state of the
receiving antennas. A Schottky diode was used as a power
detector and the output D.C. signal was fed to a micro-
ammeter for displaying the mutual coupling. A DM-5 soil
tester was fixed in the soil for measuring moisture content
where moisture content can be increased by increasing
amount of water.
The mutual coupling from only parallel polarization,
which has level about ten times higher level than the
perpendicular counterpart may provide good correlation
with moisture content but comparing with the difference
from the parallel and perpendicular ones, the difference
provides better result. Hence, we selected to find moisture
content from the measure difference of both polarizations.
Although other polynomials can provide better
approximation to the measured results, the power
approximation has sufficiently accuracy and the moisture
content can be quickly estimated. This is important for real
time moisture content measurement in the wide area.
The performance of the sensor showing the predict-
measure relation from the experiment is much better than
the measured results using only one polarization. The
accuracy is 98.6%.
4. SummaryAccording to the requirement of a sensor for measuring
moisture content of soil in a wide area, a sensor that can
provide real time results is proposed. The sensor can
measure moisture content by placing it on the surface of
soil and the good result is obtained from difference of
mutual coupling from parallel and perpendicular polarized
dipole antennas. Since the frequency at VHF band of 245
MHz can penetrate deep in the soil and a low cost device
can be obtained from this frequency band, the sensor was
designed at this frequency. The sensor is miniaturized by
designing meander line dipole antennas. The measurement
results comparing to the soil tester is pretty good. The
result is from the fixed kind of soil and the different kind
of soil is in the further study. This sensor can be useful for
irrigation system.
References[1] A.E-C Tan, S.Richards, L.Sarrabezolles, I.Platt, and
I.Woodhead, “Proximal soil moisture sensing of dairy pasture,” IEEE APS/URSI 2015, Vancouver, 2015.
[2] S. Lekshmi, D.N. Sigh, and M.S. Baghini, “A critical review of soil moisture measurement,” Measurement , vol. 54, pp.92-105, 2014.
[3] J.L. Nicole, “The input impedance of horizontal antennas above an imperfect earth,” Radio Science, Vol.15, pp.471-477, 1980.
[4] J. Mearnchu, T. Limpiti, D. Torrungrueng, P. Akkaraekthalin, and M. Krairiksh, “A handheld moisture content sensor using coupled-dipole antennas,” Latin Amer. Appl. Res., vol.40, No.3, pp.199-206, 2010.
[5] T.Limpiti and M.Krairiksh, “In Situ moisture content monitoring sensor detecting mutual coupling magnitude between parallel and perpendicuar dipole antennas,” IEEE Trans. Instrumentation and Measurement, vol.61, no.8, pp.2230-2241, Aug. 2012.
[6] C.M.K. Gardner, T.J. Dean, and J.D. Cooper, “Soil
water content measurement with a high-frequency
capacitance sensor,” J. Agric. Eng. Res., Vol. 71 ,
pp.395-403, 1998.[7] H.Nakano, H.Tagami, A.Yoshizawa, and J.Yamauchi,
“Shortening of modified dipole antennas,” IEEE Trans. Antennas and Propagation, vol.AP-32, pp.385-386, Apr.1984.
27
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Design of a Metamaterial Absorber for 24 GHz Automotive Radar System
Jinpil Tak, Eun Jeong, and Jaehoon Choi
Department of Electronics & Computer Engineering, Hanyang University, Seoul, Republic of Korea
[email protected] (corresponding author)
Ⅰ. IntroductionElectromagnetic (EM) absorbers are generally classified
into three types. The most common EM absorber is an
attenuation type which has pyramid-shaped array. It is
commonly used in an anechoic chamber. However, the
absorber is bulky, easily damaged, and expensive. Another
type is conductive loss or magnetic loss type [1]. However
it is quite expensive. The third type is resonance type. A
metamaterial (MTM) absorber is an electric resonant type
absorber, in general, which is also called frequency
selective surface (FSS) absorber. Many researchers pay
great attention to the dual band MTM absorber because it
inherently has narrow bandwidth [2].
In this paper, a bandwidth-enhanced double resonance
absorber is proposed. The proposed FSS absorber operates
at 24 GHz with dual absorption peaks for broad absorption
band.
Ⅱ. Geometry and Simulated resultsThe unit cell of MTM absorber consists of resonant
patch, ground, and FR4 substrate, and has a
pinwheel-like slot having different slot arms (long
arms and short arms) to achieve polarization
insensitivity and broad bandwidth characteristic as
shown in Figure 1(a). Figure 1(b) shows 90°
clockwise rotated unit cell. The sub-array is
composed of two unit cells and two rotated unit cells
with offset locations as shown in Figure 1(c).
Through simulation with an infinite array of sub-
array unit, the proposed absorber has two absorption
peaks, placed at 24.1GHz with 97.8% of absorptivity
and at 25.2 GHz with 74.5% of absorptivity in Figure
1(d). The proposed MTM absorber has 1.9 GHz of
full-width at half-maximum (FWHM).
Ⅲ. ConclusionIn vehicular environments, 24 GHz radar is used
for a core sensor of the safety system such as
collision avoidance or blind spot detection. The
proposed absorber can be used to false image signals
detected by an automotive radar.
(a) (b)
(c)
(d)
Figure 1. Geometry of the MTM absorber and simulated
results: (a) the unit cell, (b) 90° clockwise rotated unit cell,
(c) the sub-array unit, (d) simulated absorptivity and
reflectance characteristics (In infinite array).
References
[1] C.P. Nep, “Optimization of carbon fiber composite
for microwave absorber”, IEEE Transactions on
Electromagnetic Compatibility, vol. 46, no. 1,
pp.102-106, 2004.
[2] P. V. Tuong, J. W. Park, J. Y. Rhee, K. W. Kim, W.
H. Jang, H. Cheong, and Y. P. Lee, “Polarization-
insensitive and polarization-controlled dual-band
absorption in metamaterials,” Applied Physics Letters,
vol. 102, no. 8, 2013.
Acknowledgements
This research was supported by the Korea Ministry
of Land, Infrastructure, and Transport. It was also
supported by the Korea Agency for Infrastructure
Technology Advancement (Project No.: 15PTSI-
C054118-07)
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The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Cavity-Backed Printed-Dipole Antenna for Millimeter-Wave Applications
Ikmo Park and Son Xuat Ta
Department of Electrical and Computer Engineering
Ajou University, Suwon, Republic of Korea
Ⅰ. IntroductionIn recent years, printed antennas have attracted much
interest for millimeter-wave applications because of low
cost, ease of fabrication, wide bandwidth, and high-
efficiency operation [1], [2]. Printed T-dipole antennas fed
by integrated balun have been widely developed for
wireless communications [3]. The T-dipole antennas can
achieve wideband or multiband operations, but they have a
relatively low high-gain.
In this paper, we present a planar printed-dipole antenna
for use in millimeter-wave applications. We utilized a
cavity to improve the radiation characteristics of the
printed dipole in terms of its gain and similar beamwidths
in the E- and H-planes. The electromagnetic simulator of
CST Microwave Studio was used for this work.
Ⅱ. Antenna GeometryThe geometry of the proposed antenna, which is
composed of a cavity and an angled dipole, is shown in Fig.
1. The angled dipole, which is the primary radiation
element of, was printed on an RT/Duroid 5880 substrate
with a dielectric constant of 2.2 and a thickness of 0.254
mm. The radiator consists of two identical 45° angled arms,
with one on the top side and the other on the bottom side of
the substrate. The antenna was designed to match a
microstrip line. The antenna was fed by a microstrip line,
which transits to a parallel-plate transmission line of the
angled dipole. The cavity is divided into the front and rear
parts for ease of installation. The substrate having an
angled dipole was clamped between the two parts of the
cavity. The antenna, with a fixed-cavity aperture of 0.5• ×
0.5• , was optimized in terms of good impedance matching
and high gain at 28 GHz using CST Microwave Studio.
Aluminum cavity
Substrate
Hole for nut
x y
z
Angled dipole with feedline
Figure 1. Geometry of the antenna.
Ⅲ. ConclusionA cavity-backed printed-dipole antenna has been
described for millimeter-wave applications. The use of the
cavity enhanced the radiation characteristics of the angled
dipole antenna including the gain, back radiation, and
beamwidths in the E- and H-planes.
References
[1] Y. Suh and K. Chang, “A new millimeter-wave
printed dipole phased array antenna using microstrip-
fed coplanar stripline tee junction,” IEEE Trans.
Antennas Propag., 52(8), pp. 2019–2026, Aug. 2008.
[2] R. Alhalabi and G. Rebeiz, “High-efficiency angled-
dipole antennas for millimeter-wave phased array
applications” IEEE Trans. Antennas Propag., 56(10),
pp. 3136–3142, Oct. 2008.
[3] B. Edward and D. Rees, “A broadband printed dipole
with integrated balun,” Microw. J., pp. 339–344, May
1987.
(Invited paper)
29
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Waveguide Short-slot 2D-plane Coupler
for 2D Beam-switching Butler Matrix
Jiro Hirokawa and Dong-Hun Kim
Tokyo Institute of Technology
Ⅰ. IntroductionThe conventional short-slot one-plane coupler [1] is used
as hybrid or cross coupler in the one-dimensional beam-
switching Butler matrix [2]. This paper presents the
waveguide short-slot two-plane coupler as shown in Fig.1
for the two-dimensional beam-switching Butler matrix.
Ⅱ. StructureThe short slot two-plane coupler has 2x2 ports at each end
of the coupled region. The cross section shape of the coupled
region is changed from a rectangular. It has concaves at the
center of the top and the bottom sides and at the corners to
keep the symmetry in both the horizontal and the vertical
directions. The modes considered in the coupled region are
TE10, TE01, TE20, TM11, TE11, TM21, TE21 and TE30
modes of the corresponding multimode rectangular
waveguide. The cross section shape of the coupled region
should satisfy the following five conditions.
(1) TE10-like mode does not couple with the dominant
mode of the ports.
(2) TE21-like and TE30-like modes should be attenuated.
(3) The propagation constants of TE20-like, TM11-like
and TE11-like modes should be equal. Because TM11-like
and TE11-like modes can be dealt as one mode, it is called
as TM/TE11-like mode thereafter. (4) When the propagation constants of TE01-like, TE20-
like and TM21-like modes are 10β , 20β and 21β ,
respectively, 10 2120 2
β ββ
+= should be satisfied.
(5) TE10-like, TE20-like, TM/TE11-like and TM21
modes should have equal coupling with the dominant modes
of the ports.
The ideal operation of the hybrid is explained as follows.
For an incidence from Port 1 as an example, Ports 1-4 have
no outputs and Ports 5-8 have equal division in amplitude.
Ports 6 and 7 have 90-degree delay and Port 8 has 180-
degree delay in comparison with Port 5. The ideal operation
of the cross coupler is described as follows. For an incidence
from Port 1 as an example, only Port 8 have output and Ports
1-7 have no output in amplitude.The length l of the coupled region should satisfy
( )10 20 2 4
l πβ β− = for the hybrid and ( )10 20 2 2
l πβ β− =
for the cross coupler.
Ⅲ. ConclusionThe five conditions on the modes in the coupled region of
the short slot two-plane coupler has been shown.
The two-plane hybrid act as the 2x2-way two-
dimensional beam-switching Butler matrix. The measured
radiation patterns of the hybrid will be shown in the
workshop.
References
[1] H.J.Riblet, “The short-slot hybrid junction,” Proc. IRE,
vol. 40, no. 2, pp.180-184, Feb. 1952.
[2] J.Butler and R. Lowe, “Beam-forming matrix
simplifies design of electronically scanned antennas,”
Electron. Des., vol. 9, no. 8, pp. 170-173, Apr. 1961
Fig. 1. Waveguide Short-slot 2-plane Coupler.
(Invited paper)
30
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Millimeter-wave Reflectarray Antennas with
Dual-reflector Configurations
Ji Hwan Yoon and Young Joong Yoon
Yonsei University
Ⅰ. IntroductionReflector antennas are useful for the applications where
highly directive radiation patterns are required. A most
common type reflector antenna to achieve pencil beam is a
parabolic antenna which consists of a reflector with
paraboloidal surface and a feeder at the focal point of the
paraboloid [1]. Since the feeder has to be located at the
face of the reflector surface, the antenna suffers many
disadvantages such as large antenna height, additional loss
from long transmission line connecting the source at the
back of the reflector and the feeder at the front of the
reflector. By adopting dual-reflector configurations, such
as Cassegrain and Gregorian antennas, these disadvantages
can be avoided [2].
The reflector antennas can be designed with more
compact size by using microstrip reflectarrays [3].
Reflectarrays are flat reflector that consists of a number of
reflective elements. Each element provides required
reflection phase in order to compensate the different path
delays which are originated from substituting curved
reflector surface into flat surface.
In [4], microstrip reflectarray antennas with on-axis
dual-reflector configurations have been proposed, and the
design process of the dual-reflectarray that mimics the
equivalent Cassegrain or Gregorian antenna was described.
In this communication, the detailed design process and
analysis of the microstrip reflectarrays with on-axis dual-
reflector configurations are reviewed, and the design
results at millimeter wave band are presented.
Ⅱ. Design ProcessIn case of Cassegrain/Gregorian antenna, the sub-
reflector is designed as hyperboloidal/ellipsoidal surface
with two foci. The feeder is located at one focus (F1) and
the main-reflector is designed so that its focus overlaps
with the other focus (F2) of the hyperboloid. For replacing
the curved reflectors with microstrip reflectarrays, the
required reflection phase at a point S on the sub-
reflectarray is given as follows.
( ) πnSFSFkψsh
221
±−= (1)
( ) πnSFSFkψse
221
±+= (2)
where ψsh and ψse are the reflection phases for
hyperboloidal and ellipsoidal sub-reflectors, respectively.
For main-reflectarray, the required reflection phase is at a
point M is
( ) πnMFkψm 22 ±= (3)
An intrinsic problem of the on-axis dual-reflector
configuration is the blockage from the sub-reflector. By
adopting offset feeding configuration, it is possible to
avoid the blockage but the volume of the antenna has to be
increased. As a future work, it is desirable to adopt axially
displaced configuration to avoid blockage from the sub-
reflector. The detailed design process will be presented at
the conference.
References
[1] P. J. B. Clarricoats and G. T. Poulton, “High
efficiency microwave reflector antennas-A review,”
Proc. IEEE, vol. 65, pp. 1470–1504, Oct. 1977.
[2] P. W. Hannan, “Microwave antennas derived from
the Cassegrain telescope,” IRE Trans. Antennas
Propag., vol. AP-9, pp. 140–153, Mar. 1961.
[3] J. Huang and J. A. Encinar, Reflectarray Antennas.
Hoboken, NJ: Wiley–IEEE, 2008.
[4] J. H. Yoon, Y. J. Yoon, W. Lee, and J. So, “Axially
symmetric dual-reflectarray antennas,” Electron.
Lett., vol. 50, no. 13, pp. 908–910, Jun. 2014..
(Invited paper)
31
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Gain Improvement of Shaped-beam Reflector Using Simultaneous
Design of a Multimode Horn and Shaping Functions (Invited paper)oYoshio Inasawa, Takashi Tomura, Michio Takikawa, and Hiroaki Miyashita
Mitsubishi Electric Corporation
1. Introduction
A shaped beam reflector antenna can realize complex
coverage such as Japan area and is widely used for satellite
onboard antennas. The gain enhancement in shaped beam
antenna is desirable for the improvement of quality in the
service area. We investigate a design method to improve
the performance of shaped beam reflector and verify the
effectiveness of the design method.
2. Design Method
In general, the primary radiator and the main reflector
would be separately optimized in the design of shaped
beam reflector. We investigate a design method for the
shaped beam reflector by simultaneously optimizing the
primary radiator and shaping functions of main reflector.
The primary radiator is a multimode horn antenna defined
by a combination of waveguide modes. The mode
excitation ratio and the aperture diameter of the primary
radiator are optimized by PSO (Particle Swarm
Optimization). Shaping functions of the main reflector are
defined by the Fourier-Bessel series and the coefficients of
the shaping functions are optimized by CG (Conjugate
Gradient) method.
3. Verification
The simultaneous design method is implemented with
Japan coverage area. The frequency is in the Ku-band and
the initial surface of the main reflector is a paraboloid. For
the excitation modes of the primary radiator, three cases
are investigated: One mode (EH11), Two modes (TE11,
TM11) and Six modes (TE11, TM11, TE12, TM12, TE13,
TM13). The variables optimized by PSO are the horn
aperture diameter and the mode excitation ratio. The target
gain in the objective function is set to the same value for all
evaluation points. Here the ideal gain is 44.8 dBi for the
solid angle (1.35 deg^2) of the coverage.
The specifications of the optimized primary radiator and
the obtained minimum gain are shown in Table 1. The
result for a primary radiator of EH11 mode with typical
edge level of -20 dB is also shown for comparison. The
gain for all evaluation points is shown in Fig. 1. The gain
enhancement compared to the result of EH11 is also shown.
The EOC gain is highest in the case of 6 modes as the
primary radiator. Figure 1 shows the EOC gain is improved
at many evaluation points. It is confirmed that the
minimum EOC gain is improved by 0.3dB.
4. Conclusion
We have verified the effectiveness of the proposed
method for simultaneously optimizing the primary radiator
and the primary radiator.
Table 1. Design Results Primary radiator
EH11 EH11TE11,TM11
6 Modes
Parameters optimized by
PSO
HornDiameter
Horn Diameter
Mode Ratio
Mode Excitation
Ratio- - 1:0.49
1.00:0.61:0.40:0.42:0.38:-0.08
Horn Aperture Diameter(mm)
100 110.1 102.2 204.5
Edge Level(dB)
E-Plane -20 -28.2 -20.7 -29.4
H-Plane -20 -28.2 -23.5 -22.1Minimum Gain(dBi)
39.1 39.3 39.4 39.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
37
38
39
40
41
42
0 5 10 15 20 25 30 35 40 45 50 55 60 65
Dir
ecti
vity
enh
ance
men
t (dB
)
Dir
ecti
vity
(dB
i)
Evaluation point
EH11TE11+TM116-mode
Figure 1. Directivity and its Enhancement
(Invited paper)
32
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Introduction to 5G Communications and its Smartphone Antenna
Design Perspectives (Invited Paper)Kin-Lu Wong
Department of Electrical Engineering, National Sun Yat-sen University, Kaohsiung, Taiwan
.Ⅰ IntroductionOver the past two decades, the smartphone antenna has
evolved from the external antenna before the year 2000 to
the internal antenna or the casing-integrated antenna for
2G/3G/4G communications till now. Then, what will be the
next for the evolution of the smartphone antenna? It is
expected that the Massive MIMO antenna will be
perspective for the B4G/5G terminal device antenna. In
this talk, the visions of 5G mobile communications will
first be addressed. Promising 4-antenna, 8-antenna, and 16-
antenna MIMO arrays in the smartphone [1] and their
achievable MIMO channel capacities will be discussed.
.Ⅱ Multiple MIMO Smartphone AntennasTypical Massive MIMO systems for multi-users with
multi-antennas thereof are shown in Fig. 1.
Figure 1. Massive MIMO systems for multi-users.
It is known that for the LTE MIMO operation, the
achievable channel capacity increases with an increasing
number of the transmiting and receiving antennas therein.
However, owing to limited space inside the smartphone, it
has been a challenge in disposing multiple MIMO antennas
therein. Some promising multiple MIMO antennas
including four LTE low-band (824~960 MHz) antennas
(see Fig. 2) and 8-antenna and 16-antenna arrays in the
3.4~3.6 GHz (see Fig. 3) based on the open-slot antenna [2]
are presented. Their MIMO performance is also discussed.
Figure 2. Four LTE low-band (824~960 MHz) antennas.
8-antenna array 16-antenna array
Figure 3. 8-antenna and 16-antenna MIMO arrays.
.Ⅲ ConclusionMultiple MIMO antennas embedded in the smartphone
have been shown to be promising to implement. Enhanced
MIMO channel capacities can be obtained when multiple
MIMO antennas are applied.
References [1] K. L. Wong, J. Y. Lu, L. Y. Chen, W. Y. Li, and Y. L.
Ban, “8-antenna and 16-antenna arrays using the quad-antenna linear array as a building block for the 3.5-GHz LTE MIMO operation in the smartphone,” Microwave Opt. Technol. Lett., vol. 58, pp. 174-181, Jan. 2016.
[2] K. L. Wong and C. Y. Huang, “Triple-wideband open-slot antenna for the LTE metal-framed tablet device,” IEEE Trans. Antennas Propag., vol. 63, pp. 5966-5971, Dec. 2015.
(Invited paper)
33
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Low Profile PCB Integrated mmWave Array Antenna Solutions
for 5G Mobile Communication o Seungtae Ko, Youngju Lee, Kwanghyun Baek, Yoongun Kim and Wonbin Hong
Samsung Electronics, Mobile Communications Business
Ⅰ. Introduction Recently, 5G mobile communication service using
mmWave has been received attention worldwide for the
commercialization [1]. The phased array antenna has been
nominated as one of the candidates for reliable
communication service.
In this paper, several mmWave phased array antenna
solutions based on a low profile PCB substrate are
presented in order to obtain higher gain, polarization
robustness, and wide coverage.
Ⅱ. Guidelines Figure 1 shows Samsung’s antenna solution in a handset
for the 5G mobile communication. Two phased array antenna modules are inserted at corner of the mobile device and each one covers approximately 180 in order to remove the shadow region in the real time, as shown in Fig. 1(a). In addition, for stable performance under roll, pitch, and yaw motions between a transmitter (Tx) and a receiver (Rx), the interleaved array configuration is designed using a vertically polarized (VP) antenna and a horizontally polarized (HP) antenna, as shown in Fig. 1(b). The antenna elements are shown in Fig. 1(c) and a novel VP antenna having an extremely low profile is designed, especially. By
using proposed interleaved array, we can obtain the any polarization.
Since a small space will be allowed for the antenna module in the mobile device like Fig. 1(a), it is difficult to design the higher gain antenna. Currently, we expect that the limited number of antenna will be 8 or 16 elements. Because of this, the insufficient gain should be compensated from a base station for reliable communication. However, increasing the number of antenna in order to obtain higher gain is not a clear method, because feed line loss would be a critical problem. Samsung have researched the another solution to obtain highest gain as well as beam-steering property. Fig. 2 shows the proto-type planar lens antenna having an ultra-high gain for a base station. Although most lenses require large area, it will be suitable application in in the base station. Ⅲ. Conclusion In this paper, phased array antenna solutions are
presented for higher gain, polarization robustness, and wide coverage. Other valuable antenna designs and actual test results will be shown in presentation.
References [1] W. Hong, K. Baek, Y. Lee, Y. Kim, and S. Ko, “Study and
Prototyping of Practically Large-Scale mmWave Antenna
System for 5G Cellular Devices,” IEEE Communications
Magazine, vol. 52, no. 9, pp. 63-69, 2014.
Lens
Ant.
Fig. 2. Planar lens with 4X1 array antenna.
3mm
RFIC
(a) (b)
Horizontal Pol. Antenna Vertical Pol. Antenna (c)
Fig. 1. Phased array antenna module for 5G mobile communication, (a) antenna module in mobile device, (b) antenna module, (c) antenna elements.
(Invited paper)
34
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Multi-beam massive MIMO using analog-digitalhybrid configuration
Kentaro NishimoriFaculty of Engineering, Niigata University
Ikarashi 2-nocho 8050, Nishi-ku Niigata-shi, 950-2181 JapanEmail : [email protected]
I. INTRODUCTION
Recently, the concept of massive MIMO has been proposed,because massive MIMO realizes simple signal processing inMulti-user MIMO (MU-MIMO) transmission [1]. However,when the Channel State Information (CSI) feedback is em-ployed from the user terminals (UTs) to an access point (AP),this procedure gives a very large overhead compared with thecommunication data.
To solve this problem, an implicit beamforming methodwhich eliminates the CSI feedback was proposed [2]. How-ever, even if implicit beamforming is applied for the massiveMIMO system, the CSI estimation itself is still large overheadwhen considering the short packet communications such asWireless LAN systems [2].
In this paper, we propose analog-digital hybrid configura-tion using analog multi-beams with dielectric line array andlens and blind algorithm called Constant Modulus Algorithm(CMA) [3] which does not need the CSI estimation. Via acomputer simulation, the effectiveness of proposed configura-tion is verified.
II. PROPOSED METHOD AND CONFIGURATION
Fig.1 shows the configuration by the proposed method.In the proposed method, M orthogonal multiple beams areprepared at analog part. Fig. 2 shows an example of multi-beam patterns. The received powers for all the users aremeasured at the output of multiple beams. Selected numberof beams is less than number of users (K). The user trackingis realized by the beam selection without CSI estimation.
Key question is how to realize the hardware of multi-beamforing network in the analog part. It is well known thatbutler matrix realizes multi-beam pattern [4]. Fig.3 shows anexample which realizes multiple beams in the analog part.Because this configuration consists of dielectric line array andlens, multi-beam forming with low loss is expected. Moreover,because reduction effect in the antenna and circuit size due todielectric is obtained, this circuit can be applicable for not onlyhigh frequency band but also micro frequency bands which areused in latest wireless communication systems. In addition,feeder circuit and antenna can be combined inside one circuit,because the production is possible by an injection molding.
The MU-MIMO transmission without CSI estimation foreach user is realized by the circuit in Fig.3. However, whendirection of arrivals (DoAs) are closed among users, same
#1 #2
CMA (Digital)
#M
Beam forming network (Analog)
Down.conv.
A/D
Down.conv.
A/D
Beam selector#K#1
UT#1UT#2
BS
Fig. 1. Proposed configuration.
-90 -60 -30 0 30 60 90-40
-35
-30
-25
-20
-15
-10
-5
0
Angle [deg.]
Rel
ativ
e po
wer
[dB
]
Fig. 2. Multi-beam patterns (16-beam).
beam must be utilized for each user. Moreover, the signalsby users except intended user are actually received at sidelobe and the interference cannot be realized by only multi-beamforing network. In order to realize the perfect interferencerejection, digital beam forming (DBF) based blind adaptivearray is introduced at the output of selected multi beams. Inthis paper, constant modulus algorithm (CMA) is used as theblind adaptive algorithm [3]. The CMA works only receivedsignals and does not need CSI. In addition, CMA reduce the
(Invited paper)
35
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Dielectric rod
Dielectric lens
Dielectric lens
Dielectric rod
Fig. 3. Hardware configuration realizing multi-beam.
interference with environment where carrier and timing offsetexist. Hence, hybrid configuration with multi beams in theanalog part and blind algorithm in the digital part is suitablefor efficient transmission in massive MIMO system.
III. SINR CHARACTERISTICS BY PROPOSED METHOD
To verify the basic performance of proposed method, thecomputer simulation is carried out. When two user existin multi-path environment, the signal to interference plusnoise power ratio (SINR) is evaluated. In this simulation,single cluster model is assumed and angular spread at the BSassumed to be 10. The signal to noise power ratio (SNR) perantenna at the BS is 20 dB. The number of multi beams is 64.Least square method is adopted as the optimization algorithmregarding the CMA [3]. The propagation condition is changedand the number of trials is 10,000.
Cumulative Density Function (CDF) of SINR is plotted inFig. 4 when the difference of DoA (∆θ) is set to be 5◦ and30◦. Selected number of beams for each user is changed fromone to three. As can be seen in Fig. 4, the SINR is almost0 dB with CDF=1% when ∆θ = 5◦. On the other hand, theSINR with 2 beams is greater than 20 dB even in CDF=1%. Asshown in Fig. 4, although the SINR by only 1 beam withoutCMA is greater than 20 dB with CDF=1% when ∆θ = 30◦,the further improvement in the SINR is observed thanks to thecombination of CMA.
The SINR versus difference of DoAs between two users(∆θ) with CDF=10% is plotted in Fig. 5. Selected numberof beams for each user is changed from one to three. As canbe seen in Fig. 5, the SINR is greatly decreased when ∆θis less than 10◦. On the other hand, it is observed that theSINR is greater than 25 dB when the number of beams isgreater equal to two. Although the results are not plotted inthis figure, we confirmed that the SINR is not improved even ifthe number of beams is greater than three. From these results,it is shown that our multi-beam massive MIMO is effectivefor high transmission quality.
0 5 10 15 20 25 30 35 400
20
40
60
80
100
Output SINR [dB]
CD
F [%
]
1 beam
∆θ = 5 deg.
2 beams
∆θ = 30 deg.
K = 2, M = 64
Fig. 4. CDF of SINR (∆θ = 5, 30◦).
0
5
10
15
20
25
30
35
0 10 20 30 40 50 60
SIN
R [
dB]
Difference of DoA [deg.]
CDF = 10 %
1-beam2-beam3-beam
Fig. 5. SINR Characteristics versus DoA of interference.
IV. CONCLUSION
In this paper, we have proposed analog-digital hybridconfiguration using analog multi-beams with dielectric linearray and lens and blind algorithm called Constant ModulusAlgorithm (CMA) which does not need the CSI estimation.Via computer simulation, although the SINR is almost 0 dBwith CDF=1% when the difference of DoAs is 5◦, the SINRwith 2 beams is greater than 20 dB even in CDF=1%.
ACKNOWLEDGMENTS
The part of this work was supported by KAKENHI, Grant-in-Aid forScientific Research (C) 25420362.
REFERENCES
[1] F. Rusek, D. Persson, B. K. Lau, E. G. Larsson, T. L. Marzetta,O. Edfors, and F. Tufvesson, “Scaling Up MIMO –Opportunities andchallenges with very large MIMO–,” IEEE Signal Processing Magazine,pp.40–60, Jan. 2013.
[2] Hiraguri and K. Nishimori, ”Survey of transmission methods and ef-ficiency using MIMO technologies for wireless LAN systems (InvitedSurvey Paper),” IEICE Trans. Commun. Vol.E98-B, No.7, pp.1250-1267,July 2015.
[3] B. G. Agee, “The Least-Squares CMA: A New Technique for RapidCorrection of Constant Modulus Signals,” Proc.IEEE ICASSP, pp.953–956, 1986.
[4] S. Yamamoto, J. Hirokawa, and M. Ando, ”Single-Layer Hollow-Waveguide 8-Way Butler Matrix with Modified Phase Shifters, ” Proc.of ISAP2015, Sept. 2015.
36
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Measurement of Antenna Substrate by Collimated THz Waves Yuya Tojima, Hiroki Sudo, Takayuki Kubota, Keizo Cho, Hiroaki Nakabayashi, and Koji Suizu
Dept. of Electrical, Electronics and Computer Engineering, Chiba Institute of Technology
I. Introduction Recent explosive communication traffic growth has been
pushing the use of high frequency band for wireless communication system. Thus higher accuracy is required for antennas than those using in low frequency bands.
THz radio waves have received much attention in sensing various materials because THz waves are situated in between radio and light waves and there are many materials showing specific reflection and diffraction characteristics for the THz waves [1].
In this paper, in order to investigate detailed configuration of substrate which is popularly used for planar antennas, reflection characteristic of the substrate measured by collimated THz time-domain spectroscopy is reported.
Ⅱ. Measurement setup and results
Measurement setup Figure 1 shows a photo of the measurement setup. The
feature of the measurement system is using collimated THz waves instead of focused THz wave. This makes the measurement area and depth increase. Collimated THz pulse waves are illuminated to the planar substrate (Rogers RO4533) [2] and reflected waves are received at the receiver. The beam diameter of both TX and RX equipments is about 12mm. The thickness of the substrate under test is 1.6mm, and the dielectric constant of the substrate is 3.3@10GHz (Figure 2). Measurement results
Figure 3(a) shows the received pulse when metal plate is placed on the sample stage. Figure 3(b) shows the amplitude spectrum of the pulse. The maximum frequency of the pulse is about 2THz, and the amplitude is maximum at around 0.4THz. Figure 4 shows the reflection characteristics when the substrate is placed on the sample stage. Figure 4(b) is the case when metal plate is placed behind the substrate. The delay time of the pulse reflected at the other side of the substrate is found by placing the metal plate behind the substrate as shown in Fig. 4(b).
Fig. 1. Measurement setup. Fig. 2. Substrate under test.
Fig. 3 Received pulse reflected by metal plate
Fig. 4 Received pulse reflected by substrate Seven reflected pulses are observed thus the substrate should be composed of seven layers. Microscopic image will be presented at the conference.
Acknowledgement This work was supported by MEXT Supported Program
for the Strategic Research Foundation at Private Universities, Grand Number S1311004.
References [1] J. B. Jackson, et.al, IEEE Trans. Terahertz Science
and Technology, vol.1, no.1, pp.220-231, 2011. [2] http://www.rogerscorp.com/documents/888/acs/RO4
500-Series-Cost-Performance-Antenna-Grade-Laminates.pdf
Transmitter Receiver
Sample stage
-1.5
-1
-0.5
0
0.5
1
1.5
0 10 20 30 40 50Relative time [ps]
Ampl
itude
[V]
0
0.01
0.02
0.03
0.04
0.05
0 0.5 1 1.5 2Frequency [THz]
Ampl
itude
spe
ctru
m [V
rms]
(a) Received pulse (b) Amplitude spectrum
(a) w/o metal plate (b) with metal plate
-0.15
-0.1
-0.05
0
0.05
0.1
0.15
0 10 20 30 40 50Relative time [ps]
Ampl
itude
[V]
-0.8-0.6-0.4-0.2
00.20.40.60.8
0 10 20 30 40 50Relative time [ps]
Ampl
itude
[V]
Measurement of Antenna Substrate by Collimated THz Waves Yuya Tojima, Hiroki Sudo, Takayuki Kubota, Keizo Cho, Hiroaki Nakabayashi, and Koji Suizu
Dept. of Electrical, Electronics and Computer Engineering, Chiba Institute of Technology
I. Introduction Recent explosive communication traffic growth has been
pushing the use of high frequency band for wireless communication system. Thus higher accuracy is required for antennas than those using in low frequency bands.
THz radio waves have received much attention in sensing various materials because THz waves are situated in between radio and light waves and there are many materials showing specific reflection and diffraction characteristics for the THz waves [1].
In this paper, in order to investigate detailed configuration of substrate which is popularly used for planar antennas, reflection characteristic of the substrate measured by collimated THz time-domain spectroscopy is reported.
Ⅱ. Measurement setup and results
Measurement setup Figure 1 shows a photo of the measurement setup. The
feature of the measurement system is using collimated THz waves instead of focused THz wave. This makes the measurement area and depth increase. Collimated THz pulse waves are illuminated to the planar substrate (Rogers RO4533) [2] and reflected waves are received at the receiver. The beam diameter of both TX and RX equipments is about 12mm. The thickness of the substrate under test is 1.6mm, and the dielectric constant of the substrate is 3.3@10GHz (Figure 2). Measurement results
Figure 3(a) shows the received pulse when metal plate is placed on the sample stage. Figure 3(b) shows the amplitude spectrum of the pulse. The maximum frequency of the pulse is about 2THz, and the amplitude is maximum at around 0.4THz. Figure 4 shows the reflection characteristics when the substrate is placed on the sample stage. Figure 4(b) is the case when metal plate is placed behind the substrate. The delay time of the pulse reflected at the other side of the substrate is found by placing the metal plate behind the substrate as shown in Fig. 4(b).
Fig. 1. Measurement setup. Fig. 2. Substrate under test.
Fig. 3 Received pulse reflected by metal plate
Fig. 4 Received pulse reflected by substrate Seven reflected pulses are observed thus the substrate should be composed of seven layers. Microscopic image will be presented at the conference.
Acknowledgement This work was supported by MEXT Supported Program
for the Strategic Research Foundation at Private Universities, Grand Number S1311004.
References [1] J. B. Jackson, et.al, IEEE Trans. Terahertz Science
and Technology, vol.1, no.1, pp.220-231, 2011. [2] http://www.rogerscorp.com/documents/888/acs/RO4
500-Series-Cost-Performance-Antenna-Grade-Laminates.pdf
Transmitter Receiver
Sample stage
-1.5
-1
-0.5
0
0.5
1
1.5
0 10 20 30 40 50Relative time [ps]
Ampl
itude
[V]
0
0.01
0.02
0.03
0.04
0.05
0 0.5 1 1.5 2Frequency [THz]
Ampl
itude
spe
ctru
m [V
rms]
(a) Received pulse (b) Amplitude spectrum
(a) w/o metal plate (b) with metal plate
-0.15
-0.1
-0.05
0
0.05
0.1
0.15
0 10 20 30 40 50Relative time [ps]
Ampl
itude
[V]
-0.8-0.6-0.4-0.2
00.20.40.60.8
0 10 20 30 40 50Relative time [ps]
Ampl
itude
[V]
(Invited paper)
37
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
High-gain Multiband Spiral Antenna Design History for NLJD System
(Invited Paper) Kyeong-Sik Min
Department of Radio Communication Engineering, Korea Maritime and Ocean University727, Taejong-Ro, Youngdo-Ku, Busan, 606-791, Korea
Ⅰ. Introduction
This paper presents high-gain multiband spiral antenna
design history for NLJD (Non-Linear Junction Detector)
system. Detection of a tiny chip made by the semi-
conductor has been made possible by the development of
NLJD system [1]. The high-gain circular polarization
antenna has been mainly used for the NLJD system
application required high gain and high resolution.
The author has been proposed a high-gain spiral antenna
with a novel Archimedean spiral slit on the ground plane to
achieve circular polarization and designed a new cavity
added to the conical wall to realize the high gain. To realize
a higher gain and a better resolution, a biconvex dielectric
lens has been also designed and measured.
Ⅱ . Design HistoryFig. 1 shows design history of multiband spiral antenna.
Antenna structure design has been continuously conducted
to increase gain and to obtain good axial ratio.
(a) First model
(b) Cavity model
(c) Optimized slit on ground plane model
(d) Conical model
(e) Biconvex lens model
Fig. 1 Transition of spiral antenna structure design.
Ⅲ . ConclusionHigh-gain multiband spiral antenna design history for
NLJD system is explained in this paper. Biconvex lens
antenna model of Fig. 1 (e) was shown very high gain,
sharp beam and good axial ratio.
ACKNOWLEDGEMENT
This research was supported by Basic Science Research Program through the National Research Foundation of Korea (NRF) funded by the Ministry of Education (No. 2013R1A1A2059944.
References[1] Jae-Hwan Jeong, Kyeong-Sik Min and In-Hwan Kim,
“Design for High Gain Spiral Antenna by Added Conical Cavity Wall”, The journal of Korean Institute of Electromagnetic Engineering and Science, vol.13, no.3, pp. 165-172, Sep. 2013.
(1) Biconvex Lens (2) Expanded polystyrene support
(3) Spiral Antenna
(4) Conical wall
(5) Metal cap (Reflector)
38
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
B3 B2 B1
B4B5
A3 A2 A1
Building of measurement
50m
Tx positons
Tx antenna height :A1 – A3: 11.5 mB1 – B5: 2.5 m
Fig. 1 Tx positions and building of measurement
28.6
m
18.4 m
Mea
sure
men
t rou
te 1
3 m
6.2 m
1.6 m
XY
Balcony
Courtyard
1.6 m
Mea
sure
men
t rou
te 2
Fig. 2 Layout of building (2-8F) and measurement routes
A Study on Penetration Loss Modeling for 0.8 to 37 GHz Band T. Imai1, K. Kitao1, N. Tran1, N. Omaki1, Y. Okumura1, and K. Nishimori2
1 (NTT DOCOMO. INC.): 5G Laboratory, Yokosuka, Japan, [email protected] 2 (Niigata University): Faculty of Engineering, Niigata, Japan
1. IntroductionCurrently, the next generation (5G) mobile
communication system has been actively investigated all
over the world, in order to satisfy the future expected
requirements, such as super high bit rate. Here, one of the
approaches is to utilize the high-SHF band (over 6GHz)
and EHF band (mainly 30 – 100 GHz) [1, 2]. Up to now,
the path loss characteristics of the high-SHF and EHF
bands have been reported [2]. However, characteristics of
“Outdoor-to-Indoor (O2I) propagation” and its model have
not been clarified enough, even though we reported
measurement results in [3]. In this paper, penetration loss
property is clarified based on the measurement from 0.8 to
3.7 GHz in UMi (urban microcell) scenario, and they are
modeled.
2. MeasurementMeasurement was conducted in the campus of
Niigata University, Japan. Measurement environment is illustrated in Fig. 1 and 2. Frequencies used for measurement were 0.8, 2.2, 4.7, 8.4, 26.3 and 37.0 GHz. Six sleeve antennas for each frequency were installed on a car roof (at B1 – B6) or roof of building (at A1 –A3) as BS with transmission of CW signal simultaneously. The antenna heights were 2.5 or 11.5 m. Measurement was conducted with two receiver units of hand trucks. Each one installed three different sleeve antennas on it with the interval of 1.5 m between floor and the antennas. Measurement was repeated on the floors of 1st, 2nd, 4th, 6th and 8th (namely, 1F, 2F, 4F 6Fand 8F).
Path loss was calculated by post-processing the measured data of received power.
3. Extraction of penetration lossIn order to simplify the discussion, data of LOS
between Tx and building face are used in Analysis. The number of samples is 74,798.
In conventional model [4], O2I path loss in dB can be basically modeled by,
intwb PLPLPLPL ++= . (1)
Here, PLb is “basic path loss” which represents loss in outdoor scenario, PLtw is penetration loss into building, and PLin is loss inside. In this paper, PLb is assumed as free space loss;
( ) 4.32log20log20 33 +++= −− GHzinDoutDb fddPL , (2)
and PLin is assumed as
inDin dPL −= 25.0 . (3)
B3 B2 B1
B4B5
A3 A2 A1
Building of measurement
50m
Tx positons
Tx antenna height :A1 – A3: 11.5 mB1 – B5: 2.5 m
Fig. 1 Tx positions and building of measurement
28.6
m
18.4 m
Mea
sure
men
t rou
te 1
3 m
6.2 m
1.6 m
XY
Balcony
Courtyard
1.6 m
Mea
sure
men
t rou
te 2
Fig. 2 Layout of building (2-8F) and measurement routes
A Study on Penetration Loss Modeling for 0.8 to 37 GHz Band T. Imai1, K. Kitao1, N. Tran1, N. Omaki1, Y. Okumura1, and K. Nishimori2
1 (NTT DOCOMO. INC.): 5G Laboratory, Yokosuka, Japan, [email protected] 2 (Niigata University): Faculty of Engineering, Niigata, Japan
1. IntroductionCurrently, the next generation (5G) mobile
communication system has been actively investigated all
over the world, in order to satisfy the future expected
requirements, such as super high bit rate. Here, one of the
approaches is to utilize the high-SHF band (over 6GHz)
and EHF band (mainly 30 – 100 GHz) [1, 2]. Up to now,
the path loss characteristics of the high-SHF and EHF
bands have been reported [2]. However, characteristics of
“Outdoor-to-Indoor (O2I) propagation” and its model have
not been clarified enough, even though we reported
measurement results in [3]. In this paper, penetration loss
property is clarified based on the measurement from 0.8 to
3.7 GHz in UMi (urban microcell) scenario, and they are
modeled.
2. MeasurementMeasurement was conducted in the campus of
Niigata University, Japan. Measurement environment is illustrated in Fig. 1 and 2. Frequencies used for measurement were 0.8, 2.2, 4.7, 8.4, 26.3 and 37.0 GHz. Six sleeve antennas for each frequency were installed on a car roof (at B1 – B6) or roof of building (at A1 –A3) as BS with transmission of CW signal simultaneously. The antenna heights were 2.5 or 11.5 m. Measurement was conducted with two receiver units of hand trucks. Each one installed three different sleeve antennas on it with the interval of 1.5 m between floor and the antennas. Measurement was repeated on the floors of 1st, 2nd, 4th, 6th and 8th (namely, 1F, 2F, 4F 6Fand 8F).
Path loss was calculated by post-processing the measured data of received power.
3. Extraction of penetration lossIn order to simplify the discussion, data of LOS
between Tx and building face are used in Analysis. The number of samples is 74,798.
In conventional model [4], O2I path loss in dB can be basically modeled by,
intwb PLPLPLPL ++= . (1)
Here, PLb is “basic path loss” which represents loss in outdoor scenario, PLtw is penetration loss into building, and PLin is loss inside. In this paper, PLb is assumed as free space loss;
( ) 4.32log20log20 33 +++= −− GHzinDoutDb fddPL , (2)
and PLin is assumed as
inDin dPL −= 25.0 . (3)
(Invited paper)
39
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
BS
d2D-out
hBS
UT
d2D-in
hUT
θin
Fig. 3 Definitions of parameters
-10
0
10
20
30
40
0 15 30 45 60 75 90
Incident angle, θin [deg.]
Pen
etra
tion
loss
, PL t
w[d
B]
Measurement
Proposed model
Fig. 4 Penetration loss property
-30
-20
-10
0
10
20
30
1 10 100
Frequency, f [GHz]
Res
idua
l erro
r[dB
]
( )xy log484.1219.1 +−=
Fig. 5 Frequency dependent of residual error
Equations (2) and (3) are same as that in 3GPP_3D channel model [4], fGHz is frequency in GHz and definitions of other parameters are shown in the figure 3. The normalized measured data by calculated PLb and PLin are regarded as PLtw in this paper.
4. Analysis resultsFigure 4 shows the property of penetration loss,
PLtw, with respect to incident angle, θin. Note that all frequency data are plotted in this figure. We find that PLtw increases when θin becomes large and there are upper limit and lower limit. This means that this characteristic can be modeled as sigmoidal function. The proposed model shown in Fig. 4 is a regression result based on sigmoidal function, and it is expressed by
{ }19.4)-0.453(-exp16.8-21.98.6
intwPL
θ++= . (4)
Here, the values of 21.9 and 6.8 represent upper limit and lower limit, respectively. RMS value of residual error is 6.8 dB. And correlation coefficient between the residual error and incident angle is lower than 0.001. This means that the penetration loss, PLtw, is randomized by our proposed model with regard to the incident angle, enough. Figure 5 shows the frequency dependent of residual error in above mentioned regression analysis. From this figure, we can understand that frequency dependent of PLtw is very little.
5. ConclusionIn order to design the next generation (5G) mobile
communication system, it is necessary to clarify the propagation characteristics in high-SHF band (over 6GHz) and EHF band (mainly 30 – 100 GHz). In this paper, we evaluated the penetration loss property and modeled it based on measurement results from 0.8 to 3.7 GHz in UMi (urban microcell) scenario.
References [1] NTT DOCOMO, INC. “DOCOMO 5G White Paper,
5G Radio Access: Requirements, Concept and
Technologies,” July, 2014.
[2] METIS, Deliverable D1.4, METIS Channel Models,
Feb. 2015. (https://www.metis2020.com/.)
[3] T. Imai, et.al. “Study on Extension to Higher
Frequency Band of 3GPP Outdoor-to-Indoor Path
Loss Model,” ISAP2015, Nov. 2015.
[4] 3GPP TR 36.873 (V1.2.0), “Study on 3D channel
model for LTE,” Sep. 2013.
40
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Parameter Comparison of Standard Gain Horn Antenna at R-/S-/X-BandoJin-Seob Kang, Jeong-Hwan Kim, and Jeong-Il Park
Center for Electromagnetic Wave, Korea Research Institute of Standards and Science, Daejeon, Korea
Ⅰ. IntroductionSince 2009, the Antenna Measurement Club organized
by KRISS has performed an annual antenna measurement
comparison to support antenna manufacturers and end
users in Korea as a proficiency test program of the club [1].
In 2015, a comparison of power gains, radiation patterns,
and reflection coefficients was performed for three R-/S-
/X-band standard gain horn antennas in a joint effort
between KRISS and seven domestic participants from
private companies and public institutions.
Ⅱ. Description of the ComparisonThe traveling standards consist of three R-/S-/X-band
pyramidal standard gain horn antennas. The input port of
the traveling standards is connected to an output port (#1)
of a T-adapter, a short is connected to the other output port
(#2) of the three-port device, and the input port (#3) is used
as the input port of the horn antenna. The three-port device
with a short is selected to increase the reflection coefficient
of the antennas, which worsens the impedance match at the
input port.
In this comparison, antenna gain was measured by all the
participants using the gain comparison method, whereas
the extrapolation method was used at KRISS. Each of the
participants used a commercial vector network analyzer
and a conventional pattern measurement method in the far-
field region of the transmitting and receiving antennas.
Ⅲ. Measurement ResultsFig. 1 shows that swept-frequency power gains of most
of the participants, measured at the finite distances, are
roughly within ±0.4 dB from the far-field ones obtained by
KRISS at an infinite distance, while some of the
participants show gain fluctuations and deviations that
presumably are due to the lack of correct compensation for
the impedance mismatch at the receiving part of the gain
measurement system.
If a two-pole type antenna mast, which is suitable for
radiation pattern measurements of squinted beam antennas
such as base station antennas, is used for the radiation
pattern measurement of horn antennas, the poles can scatter
and block the incident wave radiated from the transmitting
antenna in the angular region far away from the bore-sight
direction of the receiving antenna. The result is co-
polarized E-plane radiation patterns that roughly agree with
each other in the angular region smaller than 45 degrees
from the bore-sight direction, as shown in Fig. 2.
Fig. 1. Swept-frequency power gains for R-band.
Fig. 2. E-plane radiation pattern at 2.6 GHz for S-band.
Ⅳ. ConclusionThe measurements show that the results for the power
gain, radiation pattern, and reflection coefficient of the R-
/S-/X-band antennas roughly agree with each other. Greater
accuracy is required for the impedance measurements of
the antennas and the measuring instruments for better
impedance mismatch correction at the connection between
the antennas and the measuring instruments, which will
provide better agreement in the gain results.
References
[1] J. Kang, J. Kim, and J. Park, “Comparison of
Antenna Parameters of R-/S-Band Standard Gain
Horn Antennas,” Journal of Electromagnetic
Engineering and Science, vol. 15, no. 4, pp. 231, Oct.
2015.
Parameter Comparison of Standard Gain Horn Antenna at R-/S-/X-BandoJin-Seob Kang, Jeong-Hwan Kim, and Jeong-Il Park
Center for Electromagnetic Wave, Korea Research Institute of Standards and Science, Daejeon, Korea
Ⅰ. IntroductionSince 2009, the Antenna Measurement Club organized
by KRISS has performed an annual antenna measurement
comparison to support antenna manufacturers and end
users in Korea as a proficiency test program of the club [1].
In 2015, a comparison of power gains, radiation patterns,
and reflection coefficients was performed for three R-/S-
/X-band standard gain horn antennas in a joint effort
between KRISS and seven domestic participants from
private companies and public institutions.
Ⅱ. Description of the ComparisonThe traveling standards consist of three R-/S-/X-band
pyramidal standard gain horn antennas. The input port of
the traveling standards is connected to an output port (#1)
of a T-adapter, a short is connected to the other output port
(#2) of the three-port device, and the input port (#3) is used
as the input port of the horn antenna. The three-port device
with a short is selected to increase the reflection coefficient
of the antennas, which worsens the impedance match at the
input port.
In this comparison, antenna gain was measured by all the
participants using the gain comparison method, whereas
the extrapolation method was used at KRISS. Each of the
participants used a commercial vector network analyzer
and a conventional pattern measurement method in the far-
field region of the transmitting and receiving antennas.
Ⅲ. Measurement ResultsFig. 1 shows that swept-frequency power gains of most
of the participants, measured at the finite distances, are
roughly within ±0.4 dB from the far-field ones obtained by
KRISS at an infinite distance, while some of the
participants show gain fluctuations and deviations that
presumably are due to the lack of correct compensation for
the impedance mismatch at the receiving part of the gain
measurement system.
If a two-pole type antenna mast, which is suitable for
radiation pattern measurements of squinted beam antennas
such as base station antennas, is used for the radiation
pattern measurement of horn antennas, the poles can scatter
and block the incident wave radiated from the transmitting
antenna in the angular region far away from the bore-sight
direction of the receiving antenna. The result is co-
polarized E-plane radiation patterns that roughly agree with
each other in the angular region smaller than 45 degrees
from the bore-sight direction, as shown in Fig. 2.
Fig. 1. Swept-frequency power gains for R-band.
Fig. 2. E-plane radiation pattern at 2.6 GHz for S-band.
Ⅳ. ConclusionThe measurements show that the results for the power
gain, radiation pattern, and reflection coefficient of the R-
/S-/X-band antennas roughly agree with each other. Greater
accuracy is required for the impedance measurements of
the antennas and the measuring instruments for better
impedance mismatch correction at the connection between
the antennas and the measuring instruments, which will
provide better agreement in the gain results.
References
[1] J. Kang, J. Kim, and J. Park, “Comparison of
Antenna Parameters of R-/S-Band Standard Gain
Horn Antennas,” Journal of Electromagnetic
Engineering and Science, vol. 15, no. 4, pp. 231, Oct.
2015.
(Invited paper)
41
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
°
,
ε
°
,
ε
°
,
ε
(Invited paper)
42
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
50° 65°
5 60°
φ
∞ ∞
∞∞
∞ ∞ τ
ε
τ
ε
β′β
φ′
β
β
φ
=
φ
43
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Direct Derivation of Closed-form Expression of Sommerfeld Integral for
Impedance Half-plane from Exact Image Formulation
Il-Suek Koh
Department of Electronic Engineering at Inha University
.Ⅰ IntroductionThe original derivation of the closed-form expression is
complicated of the Sommerfeld integral for an impedance
half-plane. This paper proposes a simple derivation.
.Ⅱ DerivationThe radiated field of an infinitesimal dipole can be
efficiently calculated by using the exact image
representation [1] given by
(1)
where 2 20zk k k , 0k is the free space
wavenumber, 2 2x x y y , and
0J is the Bessel function of the first kind. p is 0k or 0 /k , is the normalized impedance and
22R i z z . By substituting
sinhi z z i u , the integral in (1) can be
converted into
(2)
where 2 , 0/ 2i k p r z z and 0/ 2i k p r z z . When approaches infinite, u should approach / 2 i .
The integral path, C can be chosen to satisfy
Re cosh 0i u as shown in Figure 1.
Figure 1. Original integral path. The original integral path can be deformed into the path in
Figure 2. In Figure 1 & 2, 11 sinh /u z z
and 2 20 0/ve k p k p . By the Jordan’s
lemma, the integral on 4C becomes zero. Therefore,
2 53C C C . The integral on 5C can be
exactly evaluated as
5
cosh (1)00 2
i u
Ce du iH
where (1)0H is the Hankel function of the first kind.
Figure 2. Deformed integral path.
The integral on 1C can be represented in terms of
incomplete cylindrical function of Poisson form [2] as
1
1
cosh0 10
,2
u v i u
Ce du iE i u v .
The derived closed-form expression is identical to that in
[1].
.Ⅲ ConclusionThe closed-form expression of the Sommerfeld integral for
an impedance half-plane is directly formulated from the
exact image representation. The procedure is simple.
References
[1] I. Koh and J. Yook, “Exact Closed-Form
Expression of a Sommerfeld Integral for the
Impedance Plane Problem,” IEEE Trans. Antennas
Propag., Vol. 54, No. 9, PP 2568-2576, Sept. 2006.
[2] M. M. Agrest, M. S. Maksimov, Theory of
Incomplete Cylindrical Functions and Their
Applications, Springer-Verlag, New York, 1971.
0
00 0
1z
ik Rik z z p
z z
k eJ k e dk i e dk p k R
0
cosh
0
ik Rip z zp i u
C
e e d ie e duR
Re u
Im u
1u
/ 2
C
Im 0p Im 0p
Direct Derivation of Closed-form Expression of Sommerfeld Integral for
Impedance Half-plane from Exact Image Formulation
Il-Suek Koh
Department of Electronic Engineering at Inha University
.Ⅰ IntroductionThe original derivation of the closed-form expression is
complicated of the Sommerfeld integral for an impedance
half-plane. This paper proposes a simple derivation.
.Ⅱ DerivationThe radiated field of an infinitesimal dipole can be
efficiently calculated by using the exact image
representation [1] given by
(1)
where 2 20zk k k , 0k is the free space
wavenumber, 2 2x x y y , and
0J is the Bessel function of the first kind. p is 0k or 0 /k , is the normalized impedance and
22R i z z . By substituting
sinhi z z i u , the integral in (1) can be
converted into
(2)
where 2 , 0/ 2i k p r z z and 0/ 2i k p r z z . When approaches infinite, u should approach / 2 i .
The integral path, C can be chosen to satisfy
Re cosh 0i u as shown in Figure 1.
Figure 1. Original integral path. The original integral path can be deformed into the path in
Figure 2. In Figure 1 & 2, 11 sinh /u z z
and 2 20 0/ve k p k p . By the Jordan’s
lemma, the integral on 4C becomes zero. Therefore,
2 53C C C . The integral on 5C can be
exactly evaluated as
5
cosh (1)00 2
i u
Ce du iH
where (1)0H is the Hankel function of the first kind.
Figure 2. Deformed integral path.
The integral on 1C can be represented in terms of
incomplete cylindrical function of Poisson form [2] as
1
1
cosh0 10
,2
u v i u
Ce du iE i u v .
The derived closed-form expression is identical to that in
[1].
.Ⅲ ConclusionThe closed-form expression of the Sommerfeld integral for
an impedance half-plane is directly formulated from the
exact image representation. The procedure is simple.
References
[1] I. Koh and J. Yook, “Exact Closed-Form
Expression of a Sommerfeld Integral for the
Impedance Plane Problem,” IEEE Trans. Antennas
Propag., Vol. 54, No. 9, PP 2568-2576, Sept. 2006.
[2] M. M. Agrest, M. S. Maksimov, Theory of
Incomplete Cylindrical Functions and Their
Applications, Springer-Verlag, New York, 1971.
0
00 0
1z
ik Rik z z p
z z
k eJ k e dk i e dk p k R
0
cosh
0
ik Rip z zp i u
C
e e d ie e duR
Re u
Im u
1u
/ 2
C
Im 0p Im 0p
(Invited paper)
44
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
Optical beam scanning antenna for ultra high speed short range communication system
Hiroyuki Arai Dept. of Electrical and Computer Engineering
Yokohama National University 79-5, Tokiwadai, Hodogaya-ku, Yokohama-shi, Japan
Abstract— This paper presents an optical beam scanning antenna for ultra-high speed short range communication system. The antenna consists of switched beam waveguide and leaky antenna for high gain beam scanning. The beam direction is changed by switching feed waveguide and the beam scanning is given by sweeping the wavelength from 1500 to 1600 nm. The communication range is also discussed by this high gain transmission antenna and low gain receiving antenna with photo diode.
I. INTRODUCTION For 5G and beyond wireless communication systems, data
transmission rate is expected to be more than 100GBps [1] or 1TBps. To achieve ultra-high speed wireless data transmission, frequency band is required more than a tens of THz, which makes us to explore optical short range communication systems with small spot coverage area. A massive MIMO concept is expected to enhance channel capacity in frequency bands higher than current cellular systems, however it is necessary to use high speed digital processing with a large number of antenna elements. High speed beam scanning, equivalent to the massive MIMO, is given by conventional frequency scanned array, which is an attractive antenna in optical frequency range. Fiber optics technologies have been developed well, then the next frequency band to explore for the future system is optical range. To overcome the difficulty of large propagation loss in this frequency range, narrow beam and high gain antennas are required to be developed for small cell base station antennas.
A combination of frequency sweep and phased array technique was proposed for optical beam scanning arrays [2], however it is expected to use for sensor applications with narrow scanning range. A thermal switched phase shifter for two-dimensional (2D) beam scanner was also proposed for display devices [3], which is not for the application in communication system. As another beam scanning array, this paper presents leaky waffled waveguides fabricated on silicon wafer excited by beam switching circuits to scan in two orthogonal planes. Its beam scanning is given by frequency sweep and beam switching circuit is given by Mach-Zehnder (MZ) type optical phase shifter [4]. The next section describes the proposed antenna geometry and also discusses commination range.
II. OPTICAL BEAM SCANNING ANTENNA 2D beam scanning antenna consists of a beam switching
circuit and a leaky waveguide. The optical signal is coupled to the leaky waveguide and its radiated beam direction is controlled by sweeping the wavelength. This two stage beam control
provides 2D beam scanning antenna. The beam switched waveguide is given by MZ phase shifter and silicon waveguide. The leaky waveguide is fabricated by silicon photonics. The waveguide layer with 210nm thickness is put on a SiO2 substrate and is covered by 2m thickness protection layer. To achieve high gain antenna, a waffled waveguide is used in this paper, because its antenna gain is 10dB higher than a grating waveguide with the same antenna length. The period of cavity array determines the beam tilt angle which is adjusted by wavelength sweep. Current tunable lasers have the potential of 100nm variable range in wavelength, which gives 10 beam scanning. To extend its range, 5 leaky waveguides with different tilt angles are fed through optical switches.
To demonstrate beam scanning of proposed waffled waveguide, a prototype antenna is fabricated by silicon photonics. Waveguide length is L=2000Λ and the number of cavity along y axis is N=20, where Λ is the period of the cavity. A tunable laser changes input optical wavelength from 1500 to 1600 nm and its radiation is detected through lens and multi-mode fiber terminated with optical power meter. Measured output beam patterns in different input wavelength, which shows 10 beam scanning is given by proposed waffled waveguide.
The link budget is calculated for short range systems, assuming the proposed leaky transmission antenna and a photo diode reception with sensitive size of 300m and dark current of 100pA [5]. The communication range of 8m is given by 50dBi antenna gain for 1mW input power and by 45dBi for 5mW, which is acceptable design parameters for short range communication systems.
ACKNOWLEDGEMENT A part of this work was supported by JSPS KAKENHI Grant
Number 25420359.
REFERENCES [1] J. Medbo, el. al., “Channel Modelling for the Fifth Generation Mobile
Communications”, EuCAP 2014. [2] P. F. McManamon, et al., ”A review of phased array steering for narrow-
band electro optical systems”, Proc. IEEE, vol.97, no. 6, pp.1078-1096, Jun. 2009.
[3] Karel Van Acoleyen, et al., “Off-chip beam steering with a one-dimensional optical phased array on silicon-on –insulator”, Opt. Lett., vol. 34, no. 9,pp.1477-1479,2009.
[4] Y. Morimoto and H. Arai, "Wavelength-Insensitive MZ type Switch for Dielectric Leaky Wave Antenna for Optical Transmission,” B-1-182, IEICE Spring Conf., Mar. 2014.
[5] www.kyosemi.co.jp/sensor/nir_photodiode/kpde030
Optical beam scanning antenna for ultra high speed short range communication system
Hiroyuki Arai Dept. of Electrical and Computer Engineering
Yokohama National University 79-5, Tokiwadai, Hodogaya-ku, Yokohama-shi, Japan
Abstract— This paper presents an optical beam scanning antenna for ultra-high speed short range communication system. The antenna consists of switched beam waveguide and leaky antenna for high gain beam scanning. The beam direction is changed by switching feed waveguide and the beam scanning is given by sweeping the wavelength from 1500 to 1600 nm. The communication range is also discussed by this high gain transmission antenna and low gain receiving antenna with photo diode.
I. INTRODUCTION For 5G and beyond wireless communication systems, data
transmission rate is expected to be more than 100GBps [1] or 1TBps. To achieve ultra-high speed wireless data transmission, frequency band is required more than a tens of THz, which makes us to explore optical short range communication systems with small spot coverage area. A massive MIMO concept is expected to enhance channel capacity in frequency bands higher than current cellular systems, however it is necessary to use high speed digital processing with a large number of antenna elements. High speed beam scanning, equivalent to the massive MIMO, is given by conventional frequency scanned array, which is an attractive antenna in optical frequency range. Fiber optics technologies have been developed well, then the next frequency band to explore for the future system is optical range. To overcome the difficulty of large propagation loss in this frequency range, narrow beam and high gain antennas are required to be developed for small cell base station antennas.
A combination of frequency sweep and phased array technique was proposed for optical beam scanning arrays [2], however it is expected to use for sensor applications with narrow scanning range. A thermal switched phase shifter for two-dimensional (2D) beam scanner was also proposed for display devices [3], which is not for the application in communication system. As another beam scanning array, this paper presents leaky waffled waveguides fabricated on silicon wafer excited by beam switching circuits to scan in two orthogonal planes. Its beam scanning is given by frequency sweep and beam switching circuit is given by Mach-Zehnder (MZ) type optical phase shifter [4]. The next section describes the proposed antenna geometry and also discusses commination range.
II. OPTICAL BEAM SCANNING ANTENNA 2D beam scanning antenna consists of a beam switching
circuit and a leaky waveguide. The optical signal is coupled to the leaky waveguide and its radiated beam direction is controlled by sweeping the wavelength. This two stage beam control
provides 2D beam scanning antenna. The beam switched waveguide is given by MZ phase shifter and silicon waveguide. The leaky waveguide is fabricated by silicon photonics. The waveguide layer with 210nm thickness is put on a SiO2 substrate and is covered by 2m thickness protection layer. To achieve high gain antenna, a waffled waveguide is used in this paper, because its antenna gain is 10dB higher than a grating waveguide with the same antenna length. The period of cavity array determines the beam tilt angle which is adjusted by wavelength sweep. Current tunable lasers have the potential of 100nm variable range in wavelength, which gives 10 beam scanning. To extend its range, 5 leaky waveguides with different tilt angles are fed through optical switches.
To demonstrate beam scanning of proposed waffled waveguide, a prototype antenna is fabricated by silicon photonics. Waveguide length is L=2000Λ and the number of cavity along y axis is N=20, where Λ is the period of the cavity. A tunable laser changes input optical wavelength from 1500 to 1600 nm and its radiation is detected through lens and multi-mode fiber terminated with optical power meter. Measured output beam patterns in different input wavelength, which shows 10 beam scanning is given by proposed waffled waveguide.
The link budget is calculated for short range systems, assuming the proposed leaky transmission antenna and a photo diode reception with sensitive size of 300m and dark current of 100pA [5]. The communication range of 8m is given by 50dBi antenna gain for 1mW input power and by 45dBi for 5mW, which is acceptable design parameters for short range communication systems.
ACKNOWLEDGEMENT A part of this work was supported by JSPS KAKENHI Grant
Number 25420359.
REFERENCES [1] J. Medbo, el. al., “Channel Modelling for the Fifth Generation Mobile
Communications”, EuCAP 2014. [2] P. F. McManamon, et al., ”A review of phased array steering for narrow-
band electro optical systems”, Proc. IEEE, vol.97, no. 6, pp.1078-1096, Jun. 2009.
[3] Karel Van Acoleyen, et al., “Off-chip beam steering with a one-dimensional optical phased array on silicon-on –insulator”, Opt. Lett., vol. 34, no. 9,pp.1477-1479,2009.
[4] Y. Morimoto and H. Arai, "Wavelength-Insensitive MZ type Switch for Dielectric Leaky Wave Antenna for Optical Transmission,” B-1-182, IEICE Spring Conf., Mar. 2014.
[5] www.kyosemi.co.jp/sensor/nir_photodiode/kpde030
(Invited paper)
45
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
IDC
Feed
An Electrically Small Isotropic Antenna Using Folded Split Ring Resonator
(Invited Paper)o Joon-Hong Kim, Sangwook Nam
Department of Electrical and Computer Engineering, INMC, Seoul National University, Seoul, Koreao [email protected], [email protected]
Ⅰ. IntroductionWith rapid growth of wireless communication
technologies, electrically small antennas have received
great attention due to their compact size in the systems.
Also isotropic radiation pattern which shows full spatial
coverage is very useful for some applications, such as radio
frequency identification (RFID) tags and wireless access
points (AP) due to stable link connection [1]. In this paper,
an electrically small isotropic antenna using three
dimensional (3-D) folded split ring resonators (FSRR) is
presented.
Ⅱ. Folded Split Ring Resonator AntennaThe designed FSRR antenna is shown in Fig. 1. To
miniaturize the electrical size of the antenna, its
structure is based on split ring resonators (SRR)
which show isotropic radiation characteristics.
Further miniaturization of the antenna is
implemented with interdigital capacitors (IDC) at the
end of its arms. However, the SRR antenna has poor
radiation characteristics. Thus the folded structure is
applied to the compact SRR structure to improve its
radiation characteristics [2].
Figure 1. Folded split ring resonator antenna
The simulated reflection coefficient and radiation
patterns of designed FSRR are presented in Fig. 2
and Fig. 3 respectively.
Figure 2. Simulated reflection coefficient of FSRR
Figure 3. Simulated radiation pattern of FSRR
Ⅲ. ConclusionThe compact isotropic antenna using SRR and folded
structure is presented. The electrical size of the designed
FSRR is ka = 0.54 with 1.7 % fractional bandwidth and
gain deviation is about 3.5 dBi.
Acknowledgements
This research was supported by a grant to Bio-Mimetic
Robot Research Center Funded by Defense Acquisition
Program Administration, and by Agency for Defense
Development (UD130070ID).
References
[1] Pan, Y.M, Leung, K.W,, Kai Lu “Compact Quasi-
Isotropic Dielectric Resonator Antenna With Small
Ground Plane," IEEE Trans. Antennas Propag., vol.
62, no. 2, Feb. 2014.
[2] R. E. Colin, Antennas and Radiowave Propagation,
MaGraw-Hill, 1985.
46
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
47
Centum Hotel (20,Centum 3-ro, U-dong, Haeundae-gu, Busan)
Welcome Reception2 (Rio Karaoke, 181-141 Millak-dong, Suyeong-gu, Busan)
Welcome Reception1 (Ahn-Chae, 26, Centum 3-ro, U-dong)
Banquet (Tiffany21 Cruise, 168 Marine city 1-ro, Haeundae, Busan)
MAP
Welcome Reception 1 will be held on 6:00PM 27th January, 2016, which is located at
HAEUNDAECENTUM
HOTELAhn-Chae, 2nd floor at CENTUM SQUARE Building
Ahn-Chae, 2nd floor at CENTUM SQUARE Building in the front of HAEUNDAE CENTUM HOTEL.
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
48
Banquet: Tiffany 21 Night Cruise Homepage: http://tiffany21.co.kr
Two Hours 7:00PM-9:00PM, Address 168 Marine City 1-ro, Haeundae-gu, BusanProf. Jaehoon Choi (Hanyang University, Korea)
Introducing Busan's Tiffany 21 Cruise Boat
The largest marine tourism city in Korea, Tiffany 21 Cruise is one of Busan's special marine attractions. Tiffany 21 blends
cruise excursions with a fine dining experience, offering a perfect venue for a variety of customer-tailored events.
Haeundae Beach has a beautiful 1.8 kilometer-long coastline. Passengers can take in the views of the numerous coastal
sites, including Nurimaru APEC House on Dongbaek Island, 49th Square in Suyoung-gu Namcheon-dong, and Centum
City in U-dong. With Gwangandaegyo Bridge, the nation's largest marine bridge, spanning Haeundae Beach, Tiffany 21
will give you an unforgettable cruise experience.
Surrounded by a wall of rocks, Igidae has some gorgeous scenery, with dozens of magnificent cliffs jutting out of the open
sea around it. Designated a Natural Monument No. 22, Oryukdo Island adds to Busan's coastal scenery. Also, Taejongdae
Park has huge waves crashing against the rocky shore, and has a beautiful, calm forest as well.
When you pass over Busandaegyo Bridge, which connects the mainland with Yeongdo island, you will be greeted with other
famous touristic attractions, such as Busan International Film Festival (PIFF) Square and Jagalchi Fish Market. Jagalchi
Fish Market is the largest of fish market in Korea, known for the loud noise made by middle-aged men and women
haggling for the best bargain of fresh sea food.
Banquet: We will take a Bus on 6:00PM 28th January, 2016 in the hotel lobby of HAEUNDAE CENTUM HOTEL, and
leaving at 6:20PM, go to board on 7:00PM Tiffany 21 Night Cruise, Two Hours 7:00PM-9:00PM. We will take a Bus at
9:00PM and arriving at HAEUNDAE CENTUM HOTEL around 9:10PM.
Things to enjoy on Tiffany 21
The cruise has established itself as a popular tourist attraction in Busan, with its breathtaking views of the Busan Sea.
Providing a variety of cruise experiences and a fine seafood buffet, as well as beverages, both alcoholic and non-
alcoholic. Watch the sunrise, moon, fireworks, and other various events, including family functions, weddings and
birthday parties while drifting along the beautiful coastal waters of Busan. It's especially great for groups like families
or domestic and foreign tourists, as it offers a truly unique experience of Busan.
Banquet
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
49
Transportation
Transportation from Gimhae International Airport to Hotel
Distance from Gimhae International Airport to Haeundae Centum Hotel: 26.87 km
Taxi
Takes approximately 45 minutes.
Fare: 20,000 - 30,000 KRW
Airport Limousine
Airport Limousine buses to BEXCO and Haeundae New Town departs every 20 minutes.
Get off at Haeundae Centum Hotel bus stop after approximately 50 minutes.
Fare : Adults 7,000 KRW / Children 4,500 KRW
First and last buses depart at 06:45 and 22:00, respectively.
The schedule is subject to change and dependent upon traffic conditions.
Bus
City bus 307
Gimhae International Airport → BEXCO
Fare: 1,200KRW / Approx. 1 hour 28 min.
Subway
Airport station (Busan-Gimhae line) → Sasang Station (Line 2) → Centum City Station (Exit No.3)
Fare: 2,000 KRW/ Approx. 55 min
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
50
AWAP 2016Asian Workshop on Antennas and Propagation
MEMO
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
51
AWAP 2016Asian Workshop on Antennas and Propagation
MEMO
The 3rd AWAP2016 Centum Hotel in Busan, Korea, on January 27-29, 2016
52
AWAP 2016Asian Workshop on Antennas and Propagation
MEMO