EE426 LabMan Exp1 5 Spr2013

Middle East Technical University

Electrical & Electronics Engineering Department

EE 426 Antennas and Propagation

Laboratory Manual

Contents

SUP#1 Report Format and Remarks

SUP#2 Introduction to Antenna Experiments

EXP#1 Radiation Pattern and Gain Measurements of Antennas

in Anechoic Chamber

EXP#2 Radiation Pattern Measurements of Antennas

EXP#3 Polarization Pattern and Bandwidth Measurements of Antennas

EXP#4 Linear Antennas: Impedance & Pattern Measurements

EXP#5 Reflection Method for Gain Measurements

EXP#6 Electromagnetic Fields: Measurements and Aperture Field Shaping

EXP#7 Antenna Arrays

Experiment EE426-0 Lab Report Format and Remarks

Middle East Technical University i.1 Dept. of Electrical & Electronics Eng.

EE426AntennasandPropagationLaboratory

ReportFormatandRemarks

GeneralRemarksaboutLaboratorySessions

• Each student is viewed as a responsible professional in engineering, and thus highest ethical

standards are presumed.

• It is obligatory to prepare the preliminary work in order to start an experiment. Note that

doing the preliminary work means completing a considerable portion of it, preliminary work

with negligent and insufficient content cannot be accepted.

• Be punctual, laboratory sessions are strictly 4 hours. Together with your partner(s), you are

responsible to perform all the steps of the experiment, i.e. all the partners should be ready at

each laboratory session.

• All students are required to read the whole manual of the experiment, and are also assumed

they have done so, before attending the session. This is of great significance to successfully

complete the experiment, and to comprehend the main purpose. Remember that you are

always encouraged to ask what you want to know further or what you do not understand,

before/during/after the experiments.

• There will be a quiz at the end of each experiment and time allocated for the quiz is 10

minutes. Remember that doing the preliminary work on your own is the key point to be

ready for the quiz and have a successful laboratory session.

• Take a copy of your preliminary work before submitting it; you will need it while writing your

report.

Experiment EE426-0 Lab Report Format and Remarks

Middle East Technical University i.2 Dept. of Electrical & Electronics Eng.

Report Format

• The reports must be typed using a word-processor software.

• Do not forget to include the following in the header of the report document:

Name and number of the experiment together with the course name,

Date of the session and the date of the report,

Names, student IDs of the students.

• Please reduce paper consumption:

Do not use a cover page.

Print your reports using both sides of the papers.

• Your report will include the following parts:

Introduction: The purpose of this part is to generally summarize your past session. Here are

what you need to write:

Briefly explain the aim of the experiment.

Summarize the procedure, measurements and the methods used. Do not tabulate your

measurements here.

Results and Comments: The content is generally directed by the manual. Please note the

following:

Follow the same steps with the corresponding part of the manual.

Tabulate your results and draw necessary graphs. Also point out the results you observe

from the figures and/or tables.

Make comparisons of the results with the preliminary work where applicable.

State any results contradicting or matching your expectations. Discuss the results you

obtained (both expected and unexpected results).

Please comment about what you put forward as a result.

Conclusion: You are expected to sum up the whole idea of the experiment. In addition:

Write a general discussion of what you have learned.

Specifically point out the differences between ideal/theoretical cases and the practical

aspects, express your reasoning briefly.

• Label the figures and tables you include with an appropriate name and number.

• Please think of these reports as a team work. All group members are responsible from the

reports and are expected to participate in preparing them. This is a necessity not only to behave

ethically; but also to comprehend the whole laboratory process.

• Please keep this document at hand while writing the reports.

Experiment EE426-0 Introduction to Antenna Experiments

Middle East Technical University ii.1 Dept. of Electrical & Electronics Eng.

IntroductiontoAntennaExperiments

I.IntroductiontoComponentsandInstruments

The usage of the following equipment, components and the order of their connections are

usually common in all experiments:

RF (Radio Frequency) source, isolator, variable attenuator, slotted line associated with a VSWR

meter, cavity wavemeter, slide-screw tuner, matched termination, sliding short, directional coupler,

optionally a device under test (DUT), waveguide-to-coaxial adapter or waveguide-mount diode, AVO

meter, oscilloscope, etc…

II.Components

Although the basic principles are the same, microwave and antenna circuits may seem quite

different especially to a new microwave and antenna engineering student with respect to their lower

frequency counterparts. Therefore, for convenience, it will be appropriate to present a brief

description of microwave and antenna components often used.

RF Sources:

1.)GunnDiodeOscillator: Gunn diode is a bulk semiconductor (GaAs, InP etc.) device that exhibits

a negative resistance (as required for starting oscillations), which is used together with a

conventional oscillator circuitry or waveguide cavity. Gunn diode oscillators typically generate a

microwave output at X-band; but operation can be extended to higher frequency bands (up to 100

GHz) with proper design.

2.)SolidStateSources(FETMicrowaveOscillator): Field Effect Transistors (FETs) are used in

combination with voltage-variable capacitor diodes (VARACTORs) to construct variable-frequency

oscillators at microwave frequencies. Frequency of these oscillators can be displayed on a digital

meter, and the power output is usually adjusted manually.

Isolator: The isolator is mainly used for isolating the RF source from possible reflections that can

originate at various parts of the circuit. It basically consists of a ferrite material and a permanent



magnet arrangement, which polarizes the electromagnetic waves passing through it. It is a non-

reciprocal device which introduces little attenuation (theoretically zero) along the forward direction,

while providing high attenuation along the reverse direction.

Attenuator: It consists of a resistive sheet placed along the E-plane of a waveguide, which causes

attenuation. Unlike the isolator, the attenuator has a reciprocal nature (i.e., it is bi-directional).

Attenuators with high attenuation are sometimes used instead of isolators. In addition to fixed

attenuators, mechanical movement or rotation can be exploited to construct variable attenuators.

Variable attenuators are typically calibrated with respect to a scale, and they are usually employed to

measure power ratios.

Cavity Wavemeter: It is a hollow, adjustable, cylindrical waveguide cavity, which is connected to

the waveguide circuit in parallel through some holes on the waveguide broad wall. Under resonance

conditions, the electromagnetic energy guided in the waveguide circuit can be coupled through these

holes to the cavity wavemeter (the coupling diminishes rapidly away from the resonance). The cavity

can be made to resonate at a given operating frequency by properly adjusting its depth with a screw

mechanism, causing a part of the incident microwave power to be sucked into the cavity. This in turn

creates an abrupt power drop in the remaining parts of the waveguide circuit. The operating

frequency can be thus determined by detecting this power drop and reading the cavity resonance

frequency (which is indicated on a scale).

Slotted Line: It is a waveguidehaving a properly shaped slot cut along its longitudinal axis. The slot

permits one to insert a sampling probe into the waveguide in order to measure the E-field intensity

at a particular probe position. A carriage mechanism allows probe movement along the slotted line,

and enables one to monitor the longitudinal dependence of the field. The field intensity sampled by

the probe is subsequently detected by a diode connected to it.

Waveguide-mount Diode: The diode is placed in a post on the E-plane of the waveguide. The

rectified current through the diode is measured. Ideally, it is matched to the waveguide so there is no

reflection.

VSWR Meter: A fundamental microwave measurement tool is the VSWR meter, which will be

frequently utilized in this laboratory. Usage of this instrument might seem fairly complex to the

newcomer, and relevant complexity arises from its curious scale and deflection characteristics (more



on this later). Although this instrument cannot measure the exact voltage in the waveguide, such

accuracy is not necessary for VSWR measurements: We are primarily interested in the voltage ratio

(ratio of the maximum voltage to the minimum one) for VSWR measurements, and this relative

measurement can be conducted easily with the VSWR meter.

As illustrated in Fig.i-1, the VSWR meter is typically used in conjunction with a slotted line

followed by a square-law detector (diode). A probe (which is inserted in the slotted line) senses the E-

field along the waveguide, and delivers this excitation to the diode in the form of a voltage

( ∝ ). Provided the power level in the waveguide is low enough, the diode operates in its

square-law region, in which the diode I-V characteristics can be approximated by a quadratic relation.

This square-law I-V characteristic maps the input voltage () to its square as the diode current

( ∝ ). The VSWR meter then accepts this squared-voltage term, filters it, and provides a

deflection with its indicator (through a permanent-magnet moving-coil [PMMC or D’Arsonval]

movement arrangement). This deflection is proportional to the guided power at the probe location

owing to linearly related angular deflection () and current () in the D’Arsonval movement (i.e.,

∝ ∝ () ∝ ()). Consequently, one can monitor the voltage standing wave (VSW)

pattern along the slotted line by observing the angular deflection of the VSWR meter as a function of

probe position.

Fig.i-1. Illustration of the configuration for slotted line VSWR measurements.

In the light of the discussion above, one realizes that the VSWR meter provides a nice

visualization of microwave power variation along the length of the slotted line. For VSWR

measurements, however, one is mainly concerned with voltage quantities instead of power. It is a

trivial practice to relate the microwave power to voltage with the VSWR meter: For example, half-

scale deflection of the VSWR meter indicates 1/2 of the sensed microwave power and 1/√2 of the

guided voltage with respect to the corresponding full-scale quantities. It then makes sense to prepare

the VSWR meter scale as ∝ √Δ (instead of ∝ Δ) to facilitate voltage readings.

Here comes the part which causes the general confusion: The VSWR meter scale is actually

prepared as ∝ 1/√Δ – that is, lower VSWR meter readings are obtained for larger deflections of the

Slotted Line

Diode ∝

(High+Low

Frequencies)

LPF

(Low Freq.)

VSWR Meter

z

Probe Δθ ∝

Scale∝ 1/



indicator. Stated in other words, the VSWR meter scale reads 1/ instead of . In fact, this

is done for a good reason as we shall explain here: If one finds a voltage maximum along the slotted

line (as inferred from maximum indicator deflection) and sets this voltage level to unity following a

normalization, then the normalized guided voltage at any position of the slotted line can be

represented as ()/,. The VSWR meter reading transforms this normalized voltage to

,/(). At a voltage minimum (as inferred from minimum indicator deflection), the VSWR

meter should then read ,/, , which is the very definition of VSWR! For conventional VSWR

measurements, the full-scale deflection (at a voltage maximum) is set to unity in order for the VSWR

meter reading to directly yield the VSWR at a voltage minimum.

From a more practical perspective, a VSWR meter can be viewed as a bandpass audio amplifier

with an adjustable gain. The passband of the instrument is typically around 1 kHz with a bandwidth of

10-50 Hz. Consequently, the VSWR meter only responds to inputs having frequency content in this

band. A continuous-wave (CW) microwave source has a single spectral component at a few GHz, so

that VSWR meter cannot respond to such an input. To be able to conduct VSWR measurements, one

usually modulates the microwave source amplitude with a 1 kHz square wave. Referring back to Fig.i-

1, this amplitude modulation (AM) generates low and high frequency terms at the diode output. A

low-pass filter present at the input stage of the VSWR meter preserves the low-frequency terms, to

which the instrument can respond.

As discussed in the previous paragraphs, square-law characteristic of the diode is an important

requirement for correct VSWR measurements (indeed, proper operation of the VSWR meter strictly

relies on this assumption). For low-to-moderate microwave power, the diode behaves as a square-

law device; but it loses this characteristic (becomes linear) as the power level increases further

(higher-order Taylor polynomials would be required to approximate the exponential I-V

characteristic). Consequently, the VSWR meter no longer measures the VSWR accurately at high

microwave power levels.

IMPORTANT:

Never forget to apply sufficient attenuation to keep the crystal diode in square-law region for

accurate measurements in other experiments.

In order to improve the sensitivity of the VSWR meter, tuning is generally applied prior to the

measurements. The meter also provides fine and coarse gain levels. It can make measurement both

in normal and dB scales. Other details will be examined during the related experiments.



Network Analyzer: Network analyzers are used to test the active, passive, linear and nonlinear RF

circuits. There are two types of network analyzers: Scalar and vector network analyzer. Scalar

network analyzers measure only the amplitude of the signals while vector network analyzers

measure both the amplitude and phase. In the experiments, you will use the vector network analyzer

Agilent E5071C ENA located in the microwave laboratory. The network analyzers mainly measure the

scattering parameters (S-parameters, i.e. the ratios of the reflected and transmitted signals to the

incident signal) in a given frequency range. Vector network analyzers can convert reflection

coefficient values to impedance values on a Smith Chart. In order to remove most of the

measurement errors, network analyzer is calibrated prior to measurements. For more information,

you may read the Network Analyzer Basics application note from Agilent [1].

III. The Procedure for Measuring VSWR

First, make sure that the microwave source is amplitude modulated with a reasonable output

power, the diode operates properly (in its square-law region), and VSWR meter connects to the

diode. After accomplishing the optimal settings (i.e., the diode is in the square-law region, VSWR

meter is tuned to the modulating frequency [usually 1 kHz], and proper gain level settings are

selected); the sampling probe is moved along the slotted line until a maximum is reached in the field

intensity. This level is set to “1” on the VSWR meter scale by adjusting the fine-gain knob. Now, if the

probe is brought to a nearby voltage minimum point, the indicator points to the voltage standing

wave ratio.

IV. Practical Aspects

None of the previously mentioned components are ideal due to several factors such as

mismatch to the waveguide, unwanted lossy behavior, reflections from the junctions, and other

limited capacity etc. For instance, an isolator gives a little attenuation along the non-isolating

direction and it allows small amount of field traveling in the isolated direction, the latter resulting in

finite isolation. Another example would be the wavemeter, which has a high but a finite Q.

The probe is supposed to be inserted into the slot on the broad wall of the waveguide such that

it should not distort the wave pattern.

During the measurement of the wavelength, minimum of the VSW pattern is preferred because

they are less affected by the distortions introduced by the probe. Also the sharpness of the minima

makes the measurement more accurate.



The irregularities confronted in the standing wave pattern are due to mechanical defects of the

slotted line. A slope behavior can be observed in the pattern due to losses. A waveguide-mount diode

is almost matched to the waveguide.

Finally, the wavelength, , of a rectangular waveguide operating in the dominant TE10 mode is

given by the following expression:

1

1

1

2! (1)

where is the free space wavelength and ! is the larger inner dimension of the waveguide (It is 0.9

inch = 2.286 cm for WR-90 waveguides used in this laboratory).

V. Labsoft Initialization Instructions for Antenna Radiation Pattern

Measurements

1. Turn on your PC.

2. Log on as User.

3. Switch on the power of the Unitrain Module from the back side of the Unitrain Interface.

4. Launch the Labsoft software.

5. Register as Student, no password is required.

6. Select the Antenna Technology Course.

7. From the upper toolbox, choose Instruments → Level meter and click on the DRO button to

activate the transmitter unit. Make sure that the ON AIR light turns on. Using the same interface, set

mode to dBm.

8. After the antenna to be measured is mounted on, select Settings → Start Measurement from the

upper toolbox (or simply click on the toolbar).

9. To use cursors on your measured data, select Chart →Cursors from the upper toolbox (or simply

click on the toolbar).

10. To save your data, navigate to File → Export → Format and select either Chart as picture or

Values as text options according to your needs. Next, adjust your file save location by choosing

Destination→ File → File Name. It is strongly suggested to create your own folder and save into that

folder to avoid confusion. The use of WMF format is highly encouraged for your snapshots due to

their higher quality and lower file size.



VI. Network Analyzer 1-Port Calibration Instructions

Adjusting the Calibration Frequency Range

• Press Start (. 1 .) shown in Fig.i-2.

Enter the start frequency (. 7 .). Then press G (. 7 .) for GHz.

• Press Stop (. 2 .).

Enter the stop frequency (. 7 .). Then press G (. 7 .) for GHz.

• Press Avg (. 3 .) .

Choose Averaging ON using touch pad (. 6 .).

Press Return (. 6 .).

Choosing the Calibration Set

• Press Cal (. 4 .).

Press Cal Kit (. 6 .).

Choose 85052D (. 6 .).

Press Calibrate (. 6 .).

Choose 1-Port Cal (. 6 .).

Press Select Port (. 6 .).

Press 1 (. 6 .).

Connecting the Mechanical Loads (Matched Load, Short and Open Standards)

Connect a reflection standard to the port under calibration. Follow these guidelines for making a

proper coaxial connection:

a. Align the coaxial connectors of the DUT (the calibration standard in this case) and the

network analyzer port.

b. Push the connectors toward each other and let the surfaces of the connectors touch with

proper alignment. Turn the connector nut (not the body of the connectors themselves)

gently to begin securing the connection. Keep on rotating the nut until you form a finger-

tight connection (do not exert too much effort). This would establish a loose initial

connection.

c. Finalize the connection by using a torque wrench to apply a predetermined value of torque

to the male connector nut. While doing this, you may need to fix the body of the female

connector either by hand or via a regular wrench.



3 4

1

6

2

7

5

Fig.i-2. Front panel of the Agilent E5071C vector network analyzer.



Fig.i-3 summarizes the procedure described above. Follow the same procedure in reverse when you

break a coaxial connection. After you are finished with the calibration standard, place a dust cap on

its connector to prevent contamination of its contacting surfaces.

CAUTION:

• The calibration kit you are allowed to use is fabricated with tight mechanical/electrical

specifications, and it is an expensive piece of equipment. It requires proper handling and

connector care to maintain those specifications during its lifetime. Exercise great care

when working with the calibration kit to avoid costly repairs.

• For any coaxial connection, never rotate the bodies of the connectors themselves. Doing so

might damage the fragile inner mechanical parts, which might render the connector

useless. Instead, always turn the connector nut when making/breaking coaxial

connections.

Connect the mechanical load OPEN to the input port 1.

• Press Open on touchpad (. 6 .).

• Wait until you see tick label and hear a “beep” voice, then go on.

Connect the mechanical load SHORT to the input port 1.

• Press Short on the touchpad (. 6 .).

• Wait until you see tick label and hear a “beep” voice, then go on.

Connect the mechanical BROADBAND LOAD to the input port 1.

• Press Load on touchpad (. 6 .).

• Wait until you see tick label and hear a “beep” voice.

Now you will see “Done” label is activated in (. 6 .).

Press Done (. 6 .).

Now the network analyzer is calibrated and you can continue with your measurements.

(a) Align the connectors. (b) Make the initial connection,

tighten using the nut.

(c) Secure the connection with a

torque wrench.

Fig.i-3. The procedure for making coaxial connections.



VII. How to Recall Calibration Settings of Network Analyzer

• Press Save/Recall (. 5 .) shown in Fig.i-2.

Press Recall State (. 6 .).

Press File Dialog (. 6 .).

Select the file having an STA extension that matches the suitable frequency band.

VII. References

[1] Network Analyzer Basics, Agilent Technologies Inc., http://cp.literature.agilent.com/litweb/pdf/

5965-7917E.pdf

Experiment EE426-1 Radiation Pattern and Gain Measurements of Antennas

Middle East Technical University 1.1 Dept. of Electrical & Electronics Eng.

Experiment-1:RadiationPatternandGain

MeasurementsofAntennasinAnechoicChamber

I.Introduction

An antenna radiation pattern or antenna pattern is defined as a mathematical function or a

graphical representation of the radiation properties of the antenna as a function of space

coordinates. In most cases, the radiation pattern is determined in the far-field region and is

represented as a function of the directional coordinates. A three-dimensional pattern can be formed

by measuring series of two-dimensional patterns. However, practically for most applications, a few

plots of the patterns as a function of θ for some particular values of φ, together with a few plots as a

function of φ for some particular values of θ, give most of the useful and required information. The

pattern of a linearly polarized antenna is usually measured at the principle planes, E- and H-plane.

The E-plane is defined as “the plane containing the electric-field vector and the direction of

maximum radiation,” and the H-plane as “the plane containing the magnetic-field vector and the

direction of maximum radiation.” An illustration of principle planes for horn antenna is shown in

Figure 1.1. For this example, the x-z plane (elevation plane; φ = 0) is the principal E-plane and x-y

plane (azimuth plane, θ = π/2) is the principal H-plane. For a dipole antenna, these planes are

conventionally substituted to azimuth and elevation planes where dipole is placed along the z-axis;

since the definition for direction of maximum radiation becomes obscure, due to its cylindrically

symmetrical pattern around z-axis. Though, E-plane and H-plane would still be meaningful.

Figure 1.2 shows the setup for the radiation pattern measurements. The Antenna #2 is placed in

front of the antenna under test (AUT). In order to measure the radiation pattern of AUT, Antenna #2

can be moved on a circle of radius R, where the AUT is placed at the center of the circle. However,

this procedure causes an unnecessary usage of large space. In order to reduce the space

requirement, the Antenna #2 is placed stationary and AUT is rotated around its axis using the

rotating shaft. Note that it is not important to use AUT as a transmitting or receiving antenna as far

as the reciprocity theorem is concerned (Read pp. 144-150 in [1] for more information about

reciprocity theorem). However, in practice the power and the frequency of the transmitted

electromagnetic wave must be stable. Hence, the radiating antenna must be placed on a vibration-

free platform. That is why, Antenna #2 is chosen as a transmitting antenna. Moreover, it is

convenient to operate the antenna on the rotatable frame as a receiver, since wave generation setup

(transmitter) may be sensitive and bulky.



a

b

Figure 1.1. Principal E- and H-plane patterns for a pyramidal horn antenna, [1].

Pattern Measurements in Antenna Ranges

The testing of an antenna is carried out in antenna ranges. Basically there are two types of

antenna ranges: the reflection and free space ranges. The reflection and free space ranges can be

constructed in an indoor or an outdoor environment. In reflection ranges, the specular reflection

from the ground makes a constructive interference with the direct ray at the test antenna; while the

effect of specular reflections are tried to be reduced in free space ranges (for more information on

antenna ranges read pp. 1003-1019 in [1]). An anechoic chamber is one example of a free space

range in which the reflections from its walls are reduced by covering them with absorbing materials.

For pattern measurement purposes two antennas are required and according to the theory of

reciprocity mentioned above, it does not matter which one transmits or receives. The method of the

measurement is based on rotating the test antenna about an axis which passes through the phase

center of the antenna and lies perpendicular to the plane of measurement. The second antenna must

be placed on the same plane and usually it is placed such that, it receives the maximum field in the

direction towards the test antenna. Then by an automatic arrangement, it is possible to record the

received signal (diagram) or power (pattern) as a function of angle of rotation. A typical radiation

pattern setup is shown in Figure 1.2.



II. Preliminary Work

1. Read the sections related to radiation pattern measurements and polarization pattern

measurements in [1] (pp. 1021-1028 and 1038-1043 respectively).

2. Read the sections related to gain measurements in [1] (pp. 1028-1034).

3. Derive the formula for measuring the gain of an antenna by means of a standard-gain antenna

whose gain is characterized and known.

Hint: Use the Friis transmission equation:

=

4 (1)

where Pt and Pr the transmitted and received powers at the transmitting and receiving antennas,

respectively. Gt and Gr are the transmitting and receiving antenna gains. λ is the free space

wavelength and R is the distance between the antennas.

III. Experimental Procedure

The measurements will be performed under the guidance of the assistant.

1. Initiate the setup in Figure 1.2.

Transmitting Antenna

Receiving Antenna

To Computer

To Receiver

To Generator

Rotating shaft

AUT R Antenna2

Figure 1.2. Setup for the radiation pattern measurements.



2. Measure the radiation pattern of the following antennas at 9 GHz for E- and H-planes. In

addition, determine and record the maximum power received, note the polarizations, and define

the HPBW (half-power beamwidth):

a. Standard-gain horn (pyramidal horn with dimensions a=b=7.5 cm),

b. Pyramidal horn (a= 17.8 cm, b=15.4 cm),

c. E-plane sectoral horn (a= 2.3 cm, b= 10 cm).

Fill in the table below with your observations.

Antenna HPBW

(E-plane)

HPBW

(H-plane)

Max. Received

Power (dBm)

Gain

(dB)

Type of

Polarization

Standard Gain Horn

Pyramidal Horn

E-plane Sectoral Horn

3. Measure the cross pol. pattern of the standard-gain horn.

4. Measure the gain of the pyramidal horn (a= 17.8 cm, b=15.4 cm) at 9 GHz using the standard-

gain horn (a pyramidal horn antenna with dimensions a=b=7.5 cm).

5. Measure the gain of the E-plane sectoral horn (a= 2.3 cm, b= 10 cm) at 9 GHz using the standard-

gain horn.

NOTICE:

The patterns will be sent via e-mail by the assistants.

IV. Results and Comments

1. For each of the antennas measured,

a. Plot the E- and H-plane radiation patterns and indicate the polarization of the antenna

(linear, circular or elliptical).

Note: The values of the radiation pattern data are in dB scale. Plot the normalized

radiation patterns in rectangular coordinates.

b. Calculate and tabulate the half-power beamwidths (HPBW) both obtained from the

pattern data that have been sent by email and the ones that you have recorded during

the experimental procedure. Comment on the discrepancies between the HPBW values



that you have recorded and the ones obtained from the radiation pattern graphs. Also,

include the first null beamwidth (FNBW) values in both planes in your tabulation.

c. Compare and tabulate the directivity values obtained using Krauss' formula with the gain

values measured in the experiment. Explain the discrepancies between them.

2. Plot the normalized co-pol. pattern of standard horn in rectangular coordinates. Then plot the

cross-pol. radiation pattern of standard horn antenna on the same graph and comment.

Note: The values of the radiation pattern data are in dB scale. Note that the cross-pol. pattern

should be normalized with respect to the maximum of the co-pol. pattern.

3. Compare the antennas with each other in terms of their gains, HPBWs and FNBWs considering

their relative dimensions.

4. Write your overall comments about this experiment. Specifically, comment on the experimental

setup and summarize the main principles you learned in a few sentences.

V. References

[1] C. A. Balanis, Antenna Theory, Analysis and Design, 3rd Edition, John Wiley & Sons, NY, 2005.

[2] J. D. Kraus, Antennas, McGraw Hill, New York, 1988.

Experiment EE426-2 Radiation Pattern Measurements of Antennas


Experiment-2:RadiationPatternMeasurementsof

Antennas

I.Introduction

Two of the most important characteristics of an antenna are its directional diagram (or radiation

pattern) and its polarization pattern. The purpose of this experiment is to measure the radiation

pattern of wire-type antennas (also known as linear antennas) and printed patch antennas. We will

investigate the polarization characteristics and bandwidths of these antennas in Experiment-3.

For more information on radiation pattern measurements, refer to the Introduction section of

Experiment-1.

Near and Far-Field Properties

The space around an antenna is usually divided into three regions: (a) reactive near-field,

(b) radiating near-field (Fresnel), and (c) far-field (Fraunhofer) regions. These regions are designated

according to the general form of their electromagnetic fields.

Far-field region is defined as the region of an antenna where the angular field distribution is

essentially independent of the observation range. In other words, electric (or magnetic) fields have

the general expression =

(, ), where (, φ) is the angular field distribution and it does

not depend on range. In this region, the radial field component is negligible and the field components

are essentially transverse to radial (propagation) direction, thereby forming a spherical TEM wave.

A Brief Summary of the Antenna Types used in the Experiment

⁄ dipole: The half-wave dipole is one of the most commonly used antennas. Besides their simple

construction, another reason for its popularity is that its input impedance (for very thin dipoles)

= 73 + 42.5Ω

is very close to the 50Ω or 75Ω characteristic impedance of most transmission lines, which simplifies

its matching especially at resonance. In general, the input impedance of a dipole is a function of its

length: For the special choice of half-wavelength, the input resistance becomes equal to the radiation

resistance, and input reactance becomes slightly inductive (hence the half-wave dipole is not



resonant). The dipole reactance also depends on the dipole arm thickness as well as the gap between

those arms.

In order to perform a good impedance match to a transmission line, one must eliminate the

input reactance of the dipole. This can be done with an external matching network employed behind

the dipole. A more practical and popular means to cancel the half-wave dipole reactance (hence to

make it resonant) is to slightly trim its arm length (which makes it less inductive according to the

transmission line theory) [2].

Folded Dipole: Dipole antennas can be placed in parallel to produce more radiation: When dipoles

are in close proximity, they can create a radiated power that is greater than that of a single

dipole, where is the number of dipoles. Figure 2.1 illustrates a folded dipole for = 2, ! = " 2⁄ ,

# ≪ ", % ≪ ". The antenna is fed at the center of one dipole, which is a key point in feeding. At low

frequencies (when the antenna is electrically small), the current induced on the two arms tends to

cancel each other, and the antenna does not radiate well. At higher frequencies (particularly when

the arms are λ/2-long), however, the induced currents become in phase to reinforce their radiation.

Hence at resonance, this structure acts like two closely-spaced dipoles which double the radiated

field (and quadruple the radiated power). The folded dipole has a similar radiation pattern to that of

a standard dipole; but it radiates more efficiently with a radiation resistance of &'( = &()*+ =

73,2 ≅ 300Ω [3].

Figure 2.1. Folded dipole, [1].

Yagi-Uda Antenna: A Yagi-Uda antenna consists of a number of linear dipole elements. One of its

dipole elements is energized directly by a feed transmission line, while the others act as parasitic

radiators whose currents are induced by mutual coupling. The antenna is exclusively designed to

operate as an end-fire array, and this is accomplished by having the parasitic elements in the forward

direction act as directors while those in the rear act as reflectors.



The reflector is used to suppress the back-lobe radiation. For this purpose, the length of the

reflector is chosen slightly larger than the length of the driven element. It has been concluded

numerically and experimentally that the reflector spacing and size have negligible effects on the

forward gain, and large effects on the backward gain and input impedance.

To achieve the end-fire beam formation, the parasitic elements in the direction of the beam

(directors) are somewhat smaller in length than the driven element. Typically, the driven element is

resonant with its length slightly less than λ/2, whereas the length of the directors are between 0.4λ

and 0.45λ. The separation between the directors is typically 0.3λ to 0.4λ.

Figure 2.2. Five-element Yagi-Uda antenna.

Helix Antennas: Helix antennas consist of a single conductor (or multiple conductors) wound into a

helical shape. Although a helix can radiate in many modes, the axial mode and the normal mode are

the ones of general interest. The most commonly used axial mode provides maximum radiation along

the helix axis, which occurs when the helix circumference is about one wavelength. On the other

hand, the normal mode yields maximum radiation perpendicular to the helix axis, and it occurs when

the helix dimensions are small with respect to wavelength. Higher-order radiation modes are also

possible, for example when the helix dimensions exceed those required for the axial mode. In that

case, a conical or multi-lobed pattern will result as illustrated in Figure 2.3.



Figure 2.3. Helix geometry and three radiation modes [2].

Microstrip Patch Antennas: Microstrip patch antennas are popular among the antenna

community because they are low-profile, conformable to planar or nonplanar surfaces, mechanically

robust when mounted on a rigid body, and are simple and inexpensive to manufacture using modern

printed circuit technology. The microstrip patch antenna comprises a metallic patch of a given shape

(rectangular, circular, oval, etc.) placed on a thin dielectric substrate, and is backed by a ground

plane. This antenna is designed to have its pattern maximum normal to the patch, which is

accomplished by properly choosing the mode (field configuration under the patch). A rectangular

microstrip patch antenna, which we will use in the experiment, can be represented as an array of two

radiating narrow apertures (slots) constructed by the fringing fields (see Figure 2.4) [1].

Figure 2.4. Microstrip antenna [1].



λ/2 dipole Folded dipole Circularly polarized

patch

Linearly polarized

patch

3-element Yagi-Uda 5-element Yagi-Uda Helix antenna

Figure 2.5. Antennas that will be characterized in the experiment.


1. Read the section related to radiation pattern measurements in [1] (pp. 1021-1028).

2. Define the E- and H-planes of an antenna and show them for wire-type antennas. Can we define

E- and H-planes for circularly polarized antennas? Why?

3. For the antennas given below, plot the radiation patterns in E- and H-planes on the same graph,

determine the half power beamwidths (HPBW) and the first null beamwidths (FNBW).

a. λ/2 dipole antenna

b. λ/4 monopole on an infinite ground plane

4. Roughly plot the radiation patterns of the following antennas on the same graph, [1]:

a. Microstrip rectangular patch antenna

b. Yagi-Uda antenna

5. Make a comparison between a Yagi-Uda antenna and a simple λ/2 dipole in terms of beamwidth,

directivity, and back-lobe radiation.

6. Calculate the Rayleigh distance of the λ/2 dipole at 8.5 GHz, 9.0 GHz, 9.5 GHz. How does the

magnitude of the field change with the distance from the antenna up to and beyond the Rayleigh

distance?

REMARK:

Keep a copy of your preliminary work in order to compare the theoretical results you obtained

with measured ones in your report.




1. Initiate the computer controlled system (Lucas-Nülle antenna measurement setup) shown in

Figure 2.6 under the guidance of your assistants.

NOTE:

• During the measurements, the height of the antennas must be adjusted for proper vertical

alignment with respect to their centers.

• Make sure that the receiving and transmitting antennas are aligned in polarization

before conducting your measurements.

• Save your data in a file at each step. At the end of the experiment, remove your data files

from the hard disk of the computer after copying them onto a memory stick.

Figure 2.6. Setup for the radiation pattern measurements.

2. Measure and plot the axial amplitude variation of the field of a λ/2 dipole antenna at the

operating frequency (DRO frequency of the measurement system). To do this, first set the



spacing between the receiver and transmitter platforms to 20 cm. Measure and record the

received power. Increase the distance between the antennas in 10 cm steps up to 100 cm, and

record the received power at each step. Fill in Table 1 for your reference.

Table 1. Amplitude variation of the λ/2-dipole field measured in step 2.

fop=……..GHz Distance (cm) 20 30 40 50 60 70 80 90 100

Power Level (dBm)

3. Measure the radiation patterns in both E- and H-planes of the following antennas at the

corresponding DRO frequency. For each measurement, first observe the pattern in polar form

and save this polar plot to your folder. Then switch to the rectangular coordinates (in dB scale),

find the HPBW and FNBW in degrees using data cursors, and save this rectangular graph to your

folder (with the cursors clearly showing the HPBW value). In addition, fill in Table 2 for your

reference.

a. λ/2 dipole antenna

b. Monopole

c. Folded dipole

d. 3-element and 5-element Yagi-Uda antennas

e. RHCP rectangular patch antenna

f. Linearly polarized patch antenna

Table 2. Pattern data for the antennas in step 3.

Antenna Types HPBW(E-plane) FNBW(E-plane) HPBW(H-plane) FNBW(H-plane)

λ/2 dipole antenna

Monopole

Folded dipole

3-element Yagi-Uda

5-element Yagi-Uda

RHCP patch antenna

LP patch antenna




1. Plot the axial amplitude distribution of the λ/2-dipole antenna in step 2. Do not forget to convert

your measurements to linear scale. Compare this result with your answer to part 6 of the

preliminary work. Validate your measurement by fitting your measured data to an appropriate

function of the range.

2. For each antenna measured in the experiment,

a. Plot the E- and H-plane radiation patterns, and compare them with the ones obtained in

your preliminary work.

b. Determine and tabulate the HPBW and FNBW of each antenna. Compare the measured

HPBW and FNBW of the λ/2-dipole with the ones obtained in your preliminary work.

V. References


[2] J. L. Volakis, Antenna Engineering Handbook, 4th ed. New York: McGraw-Hill, 2007.

[3] R. Schmitt, A Handbook for wireless/RF,EMC, and High-Speed Electronics, Elsevier Sci., 2002.

Experiment EE426-3 Polarization Pattern and Bandwidth Measurements


Experiment-3:PolarizationPatternandBandwidth

MeasurementsofAntennas

I.Introduction

In this experiment, we will investigate the polarization pattern and bandwidth of the antennas

which were previously introduced in Experiment-2.

Polarization of an Antenna

Polarization of an antenna reflects the vector nature of its radiation pattern, and it is

conventionally defined through the polarization of the electromagnetic wave radiated from that

antenna (in transmit mode) at a given far-field observation point [1]. As may be remembered from

your earlier electromagnetic courses, the electric field vector of an electromagnetic wave traces a

path over time at a certain point in space, and this path defines the polarization of that wave. In

general, the mentioned path turns out to be an ellipse (elliptical polarization); but it is possible to

obtain a line or a circle under some special cases (leading to linear and circular polarizations,

respectively). Let us refresh our memory of these polarization types:

LinearPolarization: A time-harmonic wave is linearly polarized at a given point in space, if the

electric field vector at that point is always oriented along the same straight line at all times. This is

accomplished if the field vector possesses

a. Only one component, or

b. Two orthogonal components that are in- or out-of-phase (i.e., the phase difference between

these orthogonal components is , ∈ ℤ).

CircularPolarization: A time-harmonic wave is circularly polarized at a given point in space if the

electric field vector at that point traces a circle as a function of time. The circularly polarized wave

satisfies each of the following conditions:

a. The field must have two orthogonal components, and

b. These two field components must have the same magnitude, and

c. These two field components must have a phase difference of ()

, ∈ ℤ (i.e., they must

be in phase-quadrature).



If the fingers of the right hand follow the direction of the rotation of E-field vector and the

thumb points to the direction of propagation of the wave, then the wave is right hand circularly

polarized (RHCP). Conversely, the wave is left hand circularly polarized (LHCP) if you satisfy this

orientation with your left hand.

Elliptical Polarization: A time-harmonic wave is elliptically polarized if the electric field vector

traces an elliptical locus in space as a function of time. Although linear and circular polarizations are

special cases of elliptical polarization, in practice we reserve the term elliptical polarization to waves

which are not linearly or circularly polarized. The necessary and sufficient conditions for elliptically

polarized wave are as follows:

a. The field must have two orthogonal components, and

b. These two components can be of the same or different magnitude.

c. (i) If the two field components are not of the same magnitude, they can differ in phase by an

arbitrary amount (except for , since otherwise the wave becomes linearly polarized).

(ii) If the two components are of the same magnitude, they can differ in phase by an

arbitrary amount (except for

, since otherwise the wave becomes linearly or circularly

polarized).

If the fingers of the right hand follow the direction of the rotation of E-field vector and the

thumb points to the direction of propagation of the wave, then the wave is right hand elliptically

polarized (RHEP). Conversely, the wave is left hand elliptically polarized (LHEP) if you satisfy this

orientation with your left hand.

Polarization Pattern Measurements

For the polarization pattern measurements, two antennas are positioned as shown in Figure 2.7

so that their maximum radiation axes coincide with the rotation axis. The transmitting antenna is

linearly polarized. The test antenna is rotated in the plane of polarization and its received signal is

recorded. A plot of the received signal level as a function of rotation angle yields the polarization

pattern, from which polarization type and the ellipticity ratio can easily be inferred. Note that the

roles of the transmitting and receiving antennas can be interchanged in this setup owing to the

reciprocity theorem.



Bandwidth of an Antenna

IEEE defines the bandwidth as "the range of frequencies within which the performance of the

antenna, with respect to some characteristic, conforms to a specified standard." Conventionally,

mentioned “characteristics” may be the return loss level, polarization, radiation pattern constraints.

The operating band of the antenna can be defined as the frequency band around the resonance

frequency within which the return loss is less than a desired ratio, say 10 dB. Thus the 10 dB return

loss bandwidth corresponds to the frequency band over which at least nine tenth of the power is

transmitted to the antenna.


1. Read the section related to polarization pattern measurements in [1] (pp. 1038-1043).

2. Draw the polarization patterns for three types of polarizations. Define geometrically the

ellipticity ratio for each type of polarization.

3. Indicate the polarizations the following antennas:

a. λ/2 dipole antenna, b. λ/4 monopole antenna, c. helix antenna,

d. Yagi-Uda antenna, e. Microstrip rectangular patch.

4. Consider the return loss of a dipole antenna shown in Figure 3.1. For this antenna find:

a. 10 dB and 15 dB bandwidths.

b. VSWR 3:1 bandwidth.

8 8.5 9 9.5 10 10.5 11 11.5 12-35

-30

-25

-20

-15

-10

-5

0

S 11

(dB

)

Frequency (GHz) Figure 3.1. Return loss of a dipole antenna.




1. Initialize the Lucas-Nülle antenna measurement setup that you have used in Experiment-2. In

order to measure the polarization pattern of antennas, make the necessary changes to obtain

the setup shown in Figure 3.2.

Receiving Antenna (AUT)

Transmitting Antenna

Figure 3.2. Setup for polarization pattern measurements.

NOTE:

• Prior to the measurements, the receiving and transmitting antennas must be properly

aligned to have their antenna centers lying on the rotation axis.

• Save your data in a file at each step. At the end of the experiment, remove your data files

from the hard disk of the computer after copying them onto a memory stick.



2. Obtain the polarization patterns of the following antennas. Do not forget to save your pattern for

each antenna in polar coordinates (in dB scale).

a. λ/2 dipole antenna,

b. Linearly polarized patch antenna,

c. Circularly polarized patch antennas (both LHCP and RHCP ones),

d. RHCP helix antenna (measure the polarization pattern along and off the helix axis).

NOTE:

In order to continue with the following steps, first perform a calibration over 6.0-12.5 GHz

frequency interval using the instructions described in section VI (“Network Analyzer 1-Port

Calibration Instructions”) found in the introduction part of the manual.

3. Measure the return loss of the following antennas using Agilent E5071C Network Analyzer over

6.0-12.5 GHz frequency band, and analyze their frequency responses. After inspecting the

measured reflection coefficient, store it by pressing Save/RecallSnPS1P (make sure to assign

a distinct filename for each antenna). You will later use these files to determine the 10 dB/15 dB

bandwidths, the input impedance at the operating frequency, and the input impedance at the

frequency where the best match is achieved for each antenna type (see Table 1).

a. λ/2 dipole antenna,

b. Monopole antenna,

c. RHCP helix antenna,

d. Five-element Yagi-Uda antenna,

e. LP microstrip rectangular patch antenna.

Table 1. Measured bandwidths and impedances of the antennas in step 3.

Antenna Types Input impedance

at fop=……GHz

Input impedance value

for best match case (MHz) (MHz)

λ/2 dipole antenna …...……… Ω @ ……GHz ……dB BW: .…..dB BW:

monopole antenna …...……… Ω @ ……GHz ……dB BW: …...dB BW:

RHCP helix antenna …...……… Ω @ ……GHz ……dB BW: .…..dB BW:

5-element Yagi-Uda …...……… Ω @ ……GHz ……dB BW: .…..dB BW:

LP microstrip patch …...……… Ω @ ……GHz …...dB BW: .…..dB BW:

4. Measure the return loss of the printed spiral antenna in 100 MHz-14 GHz band, and store its data

in an S1P file. Using this file, you will later determine the 10 /15 dB bandwidths of this antenna.




1. Plot the polarization patterns of the antennas measured in step 2. Determine and tabulate their

ellipticity ratios, types and senses of polarizations. Show your calculation in detail.

2. For each antenna characterized in step 3, plot the measured return loss as a function of

frequency and draw the corresponding impedance pattern on a Smith Chart. Calculate the

bandwidth of each antenna, and determine the impedance at the operating frequency. Comment

on the bandwidths of the antennas. Do not forget to include your results in Table 1.

3. Comment on whether the antennas are properly designed or not by comparing the input

impedance values at the operating frequency and at the best-match frequency. Compare the

bandwidth of the spiral antenna with the bandwidths of the wire and patch antennas.

4. Write your overall comments about this experiment.

INFORMATION:

For return loss and Smith Chart plotting tasks, you may use the MATLAB routines provided on

METU-Online. You will also find a sample script utilizing those routines to get you started.

V. References


[2] J. L. Volakis, Antenna Engineering Handbook, 4th ed. New York: McGraw-Hill, 2007.

[3] R. Schmitt, A Handbook for wireless/RF,EMC, and High-Speed Electronics, Elsevier Sci., 2002.

Experiment EE426-4 Linear Antennas: Impedance & Pattern Measurements


Experiment-4:LinearAntennas:Impedance&Pattern

Measurements

I.Introduction

This experiment aims to investigate wire-type antennas in terms of their radiation pattern, input

impedance, bandwidth, and current/voltage distribution. The effects of a nearby ground plane on

these characteristics will also be studied. In particular, the following items will be covered:

• Radiation patterns of wire antennas (dipole, monopole antennas and log-periodic dipole

arrays) will be measured at various frequencies, and these patterns will be interpreted in

terms of electrical length.

• Ground plane effects will be investigated through measurements of a horizontal half-wave

dipole and a vertical monopole placed above a conducting plane (which simulates the earth).

• Current and voltage distributions on dipoles of different electrical length will be measured

with near-field probes.

• Input impedance and impedance bandwidth of wire antennas will be studied.

λ/2 Dipole: The input impedance of an infinitely thin, perfectly conducting, half-wave dipole

antenna is = 73.1 + 42.3Ω. This is a good approximation for a half-wave dipole constructed

from a wire of diameter 2 which is much smaller than its length 2ℓ (i.e., ≪ ℓ). To tune the

antenna, the half length must be shortened approximately by

Δℓ

ℓ =0.225 ℓ2

(1)

where ℓ = /4. A tuned dipole has a resistive input impedance about 70 Ω, and it is called the

resonant dipole. Voltage and current distributions on the half-wave dipole are shown in Figure 4.1.

Figure 4.1. Voltage and current distributions on a resonant dipole.

I I

I V

ℓ − Δℓ



For matching purposes, the characteristic impedance of the transmission line connected to the

feeding points must be equal to the input impedance of the antenna. Under matched conditions, the

maximum power will be radiated from the antenna. The efficiency of this antenna, like any other

wire-type one, is determined by the loss resistance of the wire. The loss resistance is usually

negligible at low frequencies, and it increases with the frequency because of the skin effect.

The reactance of the antenna is capacitive below the resonance frequency, and it becomes

inductive for operation above the resonance frequency. One may understand this input impedance

behavior as well as the current/voltage distribution trends by envisioning the half-wave dipole as an

open-circuited quarter-wave transmission line.

The radiation pattern of a half-wave dipole antenna is similar to that of a short dipole: It is a

“circle” in the azimuth plane (φ-plane) and a “figure-of-eight” in the elevation plane (θ-plane) of the

antenna.

When the half-wave dipole antenna is placed over a conducting plane (e.g., earth), its input

impedance deviates from the nominal 73.1 + 42.3Ω value. For a horizontal half-wave dipole above

a perfectly conducting plane, the input impedance increases with height from zero to a very high

value (with a resistive component of approximately 95 Ω), and the impedance oscillates about its

free-space value as the height increases further [1].

Balun: Balun structures are used at the feed of dipole antennas. The word “balun” is an

abbreviation for “balanced-to-unbalanced transformation”. A coaxial cable is an unbalanced

transmission line, because the inner and the outer conductors of coaxial cable are not interfaced to

the antenna in the same way. This is illustrated in Figure 4.2 (a): Unlike the inner conductor, the

outer coaxial conductor may carry current on both of its inside and outside surfaces. Due to these

multiple current paths, a nonzero current (spill-over current, ) flows to the ground on the outside

surface of the outer conductor, an outcome which disturbs the balance of the dipole arm currents (

versus − ).

Baluns can be used to balance inherently unbalanced systems, by canceling the spill-over

current (). The type of balun used in the experiment is shown in Figure 4.2 (b). It requires that one

end of a λ/4-section of an auxiliary coaxial line be connected to the outside shield of the main coaxial

line (node B), while the other end is connected to the dipole arm which is attached to the center

conductor (node C). The voltages at nodes A and C are nearly equal in magnitude (but are out-of-

phase), and the shield construction of both coaxial cables is identical so that their impedances to the

ground are similar. Accordingly, a net current of flows toward the node C which balances the

currents on the dipole arms ( − ) as shown in Figure 4.2 (b). One would also like to set = 0 for



proper dipole radiation characteristics: When the length of the auxiliary transmission line is λ/4,

will be zero so that the dipole arms will carry equal currents of .

(a) Unbalanced coaxial line (b) λ/4 coaxial balun

Figure 4.2. Unbalanced coaxial line and the quarter-wave balun.

Vertical Monopole: A vertical monopole antenna is an asymmetrical dipole antenna as shown in

Figure 4.3. According to the image theory, the effect of earth can be accounted for by considering a

symmetrical dipole which radiates only in upper half-space. Thus !! = "#!$%/2. The monopole

antenna can be directly fed by a coaxial cable, and it does not require a balun. The radiation pattern

is the same as that of dipole antenna in the upper half-space and zero in the lower half-space.

Figure 4.3. Vertical monopole antenna.

Log-Periodic Dipole Array: An array with a gradually expanding periodic structure has electrical

properties which also vary periodically in a manner depending on its structure. The geometry of the

antenna is chosen so that electrical properties repeat periodically with the logarithm of the

frequency. Log-periodic dipole array consists of a sequence of side-by-side parallel linear dipoles, as

shown in Figure 4.4.

I1 I2

I1-I2

I1

Antenna

I1

z

x

λ/4

Image

Antenna I1-I2 I1-I2 I1

A

λ/4 I2

I1 I1

I2

B

Shorted together

C



Figure 4.4. Log-periodic dipole array.

There are certain similarities between the log-periodic array and the Yagi-Uda array; however,

the log-periodic array operates in a much wider bandwidth. Unlike the Yagi-Uda array whose

geometric dimensions do not follow any pattern; the lengths (ℓ), spacings (&), diameters (') and

even gap spacing at dipole centers of the log-periodic array increase logarithmically as defined by the

inverse of geometric ratio τ:

ℓ(ℓ= &(&

= '('= 1)

(2)

Another parameter associated with log periodic array is spacing factor σ, and it is defined as

* &( &

2ℓ(

1 )

4 tan. (3)

By using these parameters, directivity of the log-periodic array can be found from the contours

provided in Figure 4.5. E-plane beamwidth is determined mostly by the dipole pattern and is

approximately 60°, whereas the H-plane beamwidth may be determined from the Kraus’ formula:

/ 41253

012 3 014

(3)

In (3), / stands for the directivity, and 012 and 014 represent the half power beamwidths (HPBW,

in degrees) in E- and H-planes respectively.

Input Impedance Measurements and Bandwidth of an Antenna

The reflection coefficient (Γ) is the ratio of the reflected wave phasor to the incident one at the

specified terminals. The input reflection coefficient Γ6ℓ, as shown in Figure 4.6, is easily formulated

using the feed line characteristic impedance (7) and the load (antenna) impedance (8). The

derivations are given below, refer to [2] for more information.



Figure 4.5. Computed contours of constant directivity versus σ and τ for log-periodic dipole arrays [1].

Figure 4.6. Parameter definitions for reflection coefficient calculations.

8999 =87 , Γ60 = 8

999 − 18999 1

|Γ6ℓ| <=1&6ℓ 1

<=1&6ℓ 1

Γ6ℓ Γ60>?@ℓ, A . B

For a lossless line: Γ6ℓ Γ60>?CDℓ

Return loss: RL 20 log76|Γ6ℓ|

Since the electrical length of the transmission line (J Bℓ) is a linear function of frequency and

the load might exhibit a nontrivial frequency response (i.e., 8 86K), the reflection coefficient Γ

varies with frequency in general. Variation of Γ with frequency translates to a similar variation in the

power being delivered to the load (∝ 1 |Γ|). This is especially true for narrowband networks, for

ℓ

Γ6ℓ

8 7, A

Γ8 Γ60



which a slight deviation from the matched operation frequency causes an appreciable drop in the

delivered power. In practice, it is difficult to realize perfect matching at a given operating frequency

due to losses, impedance variations and fabrication tolerances; that is the reason why engineers

define an acceptable reflection level and an associated bandwidth.

IEEE defines the bandwidth as "the range of frequencies within which the performance of the

antenna, with respect to some characteristic, conforms to a specified standard." Conventionally,

mentioned “characteristics” may be the return loss level, polarization, radiation pattern constraints.

The operating band of the antenna can be defined as the frequency band around the resonance

frequency within which the return loss is less than a desired ratio, say 10 dB. Thus the 10 dB return

loss bandwidth corresponds to the frequency band over which at least nine tenth of the power is

transmitted to the antenna.

In order to measure the input impedance of an antenna and its return loss bandwidth, a vector

network analyzer (VNA) is employed in the laboratory. This is an instrument which measures the

scattering parameters (S-parameters) of LTI microwave networks. When used for one-port

measurements, the measurement result corresponds to the reflection coefficient (also represented

with =). The instrument can also display the impedance of the tested network on a Smith Chart

over a frequency range. Prior to the measurements, the network analyzer is calibrated to shift the

reference planes to the ends of its test port cables.

Measurement of Current and Voltage (Amplitude) Distributions

The current (voltage) distribution on an antenna can be sampled by using magnetic (electric)

field probes. For current measurements, a small loop (Figure 4.7.a) is brought close to the antenna

conductor and the current induced in the loop is measured. According to the Faraday’s law, the

induced emf over the loop is proportional to the captured flux at the sampling position, which is in

turn proportional to the antenna current at that location. Consequently, the antenna current

distribution can be determined by monitoring the current induced on the loop (which is related to

the induced emf through the loop’s resistance) at several sampling positions.

Similarly, voltage distribution measurements can be made by using a small voltage sampling

probe. A small dipole or monopole (Figure 4.7.b) can be used for this purpose. The electric field of

the antenna (along the direction of the probe) induces a current on the dipole or monopole, and this

induced current is approximately proportional to the antenna voltage at that particular sampling

position.

In order to avoid disturbing the near-field of the test antenna (hence its current and voltage

distributions), the current/voltage sampling probes must be small with respect to the wavelength.



During the experiment, we will use a small loop antenna and a small monopole as illustrated in

Figure 4.7.

(a) (b)

Figure 4.7. (a) Shielded current sampling loop (magnetic field probe), (b) voltage sampling monopole (electric

field probe).


1. Read the sections related to the impedance measurements (pp. 1036-1038) and current

measurements (pp. 1038) in [1].

2. Indicate the polarization of a dipole antenna. For the antennas described below, draw the

radiation patterns in E- and H-planes, and determine the half-power and first-null beamwidths

(HPBW, FNBW) at each provided frequency:

a. λ/2 dipole antenna of 8.4 cm length at the resonance frequency,

b. 8.4 cm-long dipole antenna at 1.2 GHz, 1.5 GHz, 1.8 GHz,

c. 35 cm-long dipole antenna at 1.285 GHz and 1.714 GHz.

You may check your patterns with the dipole patterns provided in [1] for different electrical

lengths.

3. Calculate the resonant length of a half-wave dipole antenna made from a round copper wire of

radius = 0.3MN and operating at a frequency of 1 GHz.

4. Plot the input resistance as a function of the height (O) for the horizontal half-wave dipole in

Figure 4.8. What is the input resistance at O λ/2? Hint: See pp. 204 in [1]).



Figure 4.8. Dipole antenna placed horizontally above an infinite and perfectly conducting ground plane.

5. Calculate and plot the elevation-plane radiation patterns of the horizontal dipole in Figure 4.8 for

ℎ = 0.5λ, ℎ = 0.75λ and at ℎ = λ.

6. Calculate the input impedance of a quarter-wave vertical monopole at resonance. Plot its

radiation patterns in elevation and azimuthal planes.

7. Suppose in Figure 4.9 that the input impedance seen at the feed terminals (b-b’) is determined

as 20 − 30Ω. What is the impedance seen at the antenna terminals (a-a’)?

Hint: Refer to the “Input Impedance Measurements” section in the Introduction part.

Figure 4.9. Configuration for Q7 of preliminary work.

REMARK:

• Keep a copy of your preliminary work in order to compare the theoretical results you

obtained with measured ones in your report.

• Bring an empty 3.5" floppy disk with you.

x

h

λ/2 dipole antenna

ℓ = /8

Antenna 7 = 50Ω, A = B

a’

a

b’

b




Radiation Pattern Measurements

Use the Feedback Antenna Measurement setup.

NOTE:

• From time to time, the rotating platform of the setup does not rewind itself after pattern

measurements. In order not the damage the antenna under test (AUT) and the coaxial

cable attached to it, make sure to check this cable (unwind manually if necessary) after

you are done with a pattern measurement.

• The patterns are saved as a PPD file on the hard drive with an automatically generated

filename. The filename is based on the system clock and is of DDMMHHmm form where D:

day, M: month, H: hour, m: minute. In order to prevent file overwrite issues, make sure you

wait at least one minute from one measurement to the next.

1. Measure the radiation patterns of the following antennas at the specified frequencies. Make the

necessary arrangements to obtain the plots for both E and H planes for linearly polarized

antennas. It is sufficient to measure H-Plane pattern of one of the dipole antennas. Be sure that

for each case (E and H-plane measurements) the receiving and transmitting antennas are in the

same polarization.

a. Dipole antennas:

i. 8.4 cm long dipole antenna at the resonance frequency (as λ/2 dipole)

calculated in the preliminary work.

ii. 8.4 cm-long dipole antenna at frequencies 1.2 GHz and 1.5 GHz.

iii. 35 cm-long dipole antenna at frequencies 1.285 GHz and 1.714 GHz.

b. Log-periodic array:

i. Measure radiation patterns at frequencies, 1.2 GHz, 1.5 GHz, and 1.8 GHz.

Determine HPBWs.

ii. Measure the length of the dipoles and spacings between them. Calculate the

directivity using measured HPBW using the expressions given in Part I-

Introduction.

2. Measure the radiation patterns (in both planes) of the horizontal λ/2 dipole antenna at a

distance h=0.5λ, 0.75λ and λ from the ground plane at 1.40 GHz.

3. Measure the radiation patterns of the λ/4 monopole antenna at 1.71 GHz and at a couple of

different frequencies around this frequency to determine the radiation bandwidth of the

antenna.



Current/Voltage Distribution Measurements

4. By using magnetic and electric field probes, measure and roughly plot the voltage and current

distributions on

a. 8.4 cm-long half-wave dipole,

b. 35 cm-long dipole at frequencies 1.285 GHz and 1.714 GHz.

Impedance Measurements

NOTE: In order to perform the following steps, you need to recall the calibration of the network

analyzer for 1.0-3.0 GHz frequency band. Refer to the instructions described in section VII (“How to

Recall Calibration Settings of Network Analyzer”) found in the introduction part of the manual.

Save your data in S1P format for each measurement configuration and do not forget to grab your

S-parameter files from the network analyzer.

5. a. For the horizontal λ/2 dipole antenna (without any ground plane), measure the return loss

and determine

i. 15 dB return loss bandwidth,

ii. VSWR=2 bandwidth.

b. Determine the complex input impedance of the same dipole seen from its antenna terminals

at the resonance frequency.

Hint: Measure the length of the transmission line from the connector to the input terminals of the

dipole and calculate the phase delay through that line section (the coaxial line is filled with a

dielectric having RS = 2.2). Then, use your reasoning in question 7 of the preliminary work.

6. Set up the circuit shown in Figure 4.10. Repeat step 5 for h =0.5λ and 0.75λ.

Figure 4.10. Experimental circuit diagram.

Antenna

Network analyzer

Coaxial line

Image plane

Open circuited

stub h



7. Measure the return loss of the log-periodic array. Determine the resonance frequency and the

10 dB bandwidth of the antenna.

8. Measure the return loss of the quarter-wave monopole antenna. Determine the resonance

frequency and the 10 dB bandwidth of the antenna.


1. Plot the radiation patterns of dipole and monopole antennas (in both planes) for all cases, and

tabulate their HPBW and FNBW values. Compare the measured radiation patterns with the ones

calculated in the preliminary work and comment.

2. Comment on the frequency dependence of the dipole radiation patterns observed in step 1.a.(ii)-

(iii) of the experimental procedure.

3. For the log-periodic antenna

a. Plot the E- and H-plane radiation patterns and determine the HPBW. Comment on HPBW

variation with frequency.

b. Compare the directivity calculated using Kraus’ formula with the one obtained from the

curves provided in Figure 4.5. Comment on the discrepancies.

4. Plot the voltage and current distributions on the dipole antennas and compare them with

theoretical expectations.

5. Plot the return loss of each antenna you have measured in steps 5-8 as a function of frequency,

and determine their 10-15 dB return loss and VSWR=2 bandwidths. Comment on the results.

6. Plot the complex reflection coefficient of each antenna on a Smith chart, and calculate the input

impedance at the respective resonant frequencies.

7. For the λ/2 dipole (with and without the ground plane), calculate the impedance seen at the

antenna terminals of the dipole (show the calculation you made in step 5-6 of the experimental

procedure).

8. Comment on the impedance bandwidth and pattern bandwidth of the monopole antenna.

INFORMATION:

For return loss and Smith Chart plotting tasks, you may use the MATLAB routines provided on

METU-Online. You will also find a sample script utilizing those routines to get you started.



V. References

[1] C. A. Balanis, Antenna Theory, Analysis and Design, 3rd

ed., John Wiley & Sons, NY, 2005.

[2] D. M. Pozar, Microwave Engineering, 3rd

ed. Hoboken, NJ: Wiley, 2005.

[3] E. C. Jordan and K. G. Balmain, Electromagnetic Waves and Radiating Systems, Prentice Hall, 2nd

ed., 1968.

[4] H. Jasik, Antenna Engineering Handbook, McGraw-Hill, 1961.

Experiment EE426-5 Reflection Method for Gain Measurements


Experiment-5:ReflectionMethodforGain

Measurements

I.Introduction

Gain is one of the most important performance parameters of an antenna. Gain includes the

efficiency of an antenna in addition to its directivity. Gain of an antenna is defined as follows:

= 4RadiationIntensityTotalInputPower = 4(, ) (1)

It is conventional to specify the gain of an antenna in dB scale, which is expressed as

!" = 10 log&' (2)

The term relative gain is sometimes used explicitly to define the gain with respect to an

arbitrary reference antenna. In general, the reference antenna is a lossless isotropic radiator whose

gain is unity in all directions (in fact, (1) is based on this premise). The reference antenna can also be

any other radiator such as a dipole or a horn. A suffix appended to “dB” emphasizes this relative

definition of gain: Gain specified in dBi uses an isotropic radiator as the reference antenna, whereas

the gain specified in dBd specifies the gain relative to a dipole antenna, (1 dB=1 dBi, 1 dBd=2.15 dBi).

There are a number of techniques that can be employed to measure the gain of an antenna

[1, 2]. These methods are based on the Friis transmission equation which assumes that the

measurement system employs two antennas. The antennas are separated by a distance

greater than ()*+,-./ to satisfy the far-field criterion of each antenna.

In this experiment, the reflection method is utilized to determine the gain of an antenna [1]. In

this method, the antenna whose gain is to be determined is placed at a certain distance from an

image plane. Same antenna is used for both transmission and reception. The transmitted wave is

reflected from the image plane and then received by the antenna. These transmitted and received

waves produce a standing wave in the waveguide system. Gain of the antenna can be calculated

from the knowledge of the voltage standing wave ratio (VSWR) on the waveguide, distance between

the antenna and the image plane ((), and the frequency of operation (0). By increasing and

decreasing ( by an amount of 1/4, the phase of the incoming signal can be reversed with negligible

effect on its amplitude. Then by taking the arithmetic mean of the gains computed for these two (

values, most of the errors caused by impedance mismatches (between the antenna and the line) and

multiple reflections (between antenna and image plane) can be eliminated.



In this experiment we will measure the gains of two pyramidal horn antennas, whose

dimensions are parametrized as 3, 4 and ℎ& in Figure 5.1. More information on horn antennas is

available in [2].

w

a

b

h

h1

(a) (b)

Figure 5.1. Dimensions of the horn antenna: (a) Perspective view. (b) Side view.


1. Read the sections related to the gain measurements in [2] (pp. 1028-1035). Also get familiar

with the VSWR meter and VSWR measurements by studying the relevant sections in the

Introduction part of this manual.

2. Considering the reflection method and the associated setup in Figure 5.2, derive the gain of

the antenna in terms of wavelength 1, distance between the antenna and image plane ((), and

measured VSWR (which is equal to (1 + |Γ|)/(1 − |Γ|) , where Γ is the reflection coefficient).

3. What is the Rayleigh distance for the pyramidal horn in Figure 5.1 with a=b=7.5 cm at 8.2 GHz,

9 GHz and 9.8 GHz? For this horn antenna, calculate the minimum distance between the image

plane and the antenna for accurate measurement at 9 GHz. What would be the Rayleigh

distance if we had a=17.6 cm, b=15.4 cm at 9 GHz?

4. Calculate the gain of the horn antenna whose dimensions are specified as

a. a=b=7.5 cm, h1=14 cm at 8.2 GHz, 9 GHz, and 9.8 GHz.

b. a=17.6 cm, b=15.4 cm, h1=19.1 cm at 9 GHz.

Use the following approximate gain formula for the pyramidal horn antennas:

!" ≅ 10 × <1.008 + log&' 341?@ − ABC(D) + BE(F)G (3)

Hint: Standard WR-90 waveguide dimensions are w=2.286 cm and h=1.016 cm. In order to

calculate BC(D) and BE(F), refer to the plot given on pp. 777 in [2]. The variables s and t are

defined as

D = 4(4 − ℎ)81ℎ& ,F = 3(3 − H)

81ℎ& (4)



5. Show for a pyramidal horn antenna that, the area of the image plane must satisfy I ≥ KLML'.N&*O in

order for the main beam of the horn to be completely intercepted by the image plane. Assume

that the antenna is lossless (i.e., Gain=Directivity). Calculate and tabulate the minimum required

area of the image plane for the following horn antennas:

a. a=b=7.5 cm at 8.2 GHz, 9 GHz and 9.8 GHz,

b. a=17.6 cm b=15.4 cm at 9 GHz.

6. For the reflection method used in this experiment, the reciprocal of the reflection coefficient is

expected to be a linear function of R; however, in practice there will be a ripple superimposed

on this linear curve. Explain the reason for this ripple and suggest a simple way to eliminate the

errors in measurements.

REMARK:

Keep a copy of your preliminary work in order to compare the theoretical results you obtained

with measured ones in your report.


1. Set up the circuit whose block diagram is given in Figure 5.2. Make sure that far-field criterion is

satisfied for the subsequent measurements.

Figure 5.2. Experimental setup for the gain measurement.

2. Set f=9.8 GHz. Use the horn antenna with dimensions a=b=7.5 cm.

3. Remove the image plane and adjust the tuner screw so that the power at the coupled port of

directional coupler becomes minimum (i.e., match the antenna to the feed). Also check the VSW

pattern and the value of VSWR at this state. List your screw positions and measured VSWR in

Table 1.

Gunn

oscilator

Isolator

Tuner Horn

Antenna

R

Image

plane

VSWR Meter

Slotted Line

Matched load

Directional

Coupler



IMPORTANT:

Obtaining reliable gain values in subsequent steps strongly depends on your effort to match the

antenna at this step. In particular, try to keep VSWR below 1.1 and reduce it further if possible.

Table 1. Experimental findings for step 3.

f=9.8 GHz Horizontal screw

tuner position (mm)

Vertical screw

tuner position

(mm)

VSWR

(best match)

4. a. Place the image plane at an appropriate distance R from the antenna, and measure the VSWR

using the slotted line. Calculate the gain using the formula you have derived in your preliminary

work.

b. Move the image plane by an amount λ/4 toward or away (either will work) from the horn

antenna, and measure the gain again. Then take the arithmetic mean of the calculated gain

values.

c. Repeat steps (a)-(b) for at least three other R values. Fill in Table 2 for your reference.


f=9.8GHz R (cm) VSWR G (linear) G (linear), avg. G (dB) G (dB),

prelim. work

5. Set f=8.2 GHz and repeat steps 3-4. Fill in Table 3 for your reference.


f=8.2GHz Horizontal screw

tuner position (mm)

Vertical screw

tuner position (mm) VSWR (best match)



f=8.2GHz R (cm) VSWR G (linear) G (linear) ,avg. G (dB) G (dB), prelim. work

6. Set f=9.0 GHz and repeat steps 3-4. Fill in Table 4 for your reference.


f=9 GHz Horizontal screw

tuner position (mm)

Vertical screw


f=9GHz R (cm) VSWR G (linear) G (linear) ,avg. G (dB) G (dB), prelim. work

7. Replace the standard gain horn with the pyramidal one having a=17.6 cm, b=15.4 cm. Set

f=9.0 GHz and repeat steps 3-4. Fill in Table 5 for your reference.


f=9 GHz Horizontal screw

tuner position (mm)

Vertical screw




f=9GHz R (cm) VSWR G (linear) G (linear) ,avg. G (dB) G (dB), prelim. work


1. Tabulate the measured VSWR and gain values for both horn antennas at each measurement

frequency (i.e., Tables 1-5). Do not forget to include the calculated gain values in your

preliminary work.

2. Plot 1/|Γ(ℓ)| as a function of R for each antenna, at each respective measurement frequency.

From these plots, extract your experimental gain values. Hint: Notice that the slope of the line

best fitting your data is a measure of gain.

3. Compare the measured gain values with the ones calculated in your preliminary work. Comment

on the results. Calculate the aperture efficiency of horn antennas. Note that the relation

between gain () and effective aperture (I-) is given by

= 41? I- , I- =Q*RIR

where Q*R is the aperture efficiency and IR is the physical aperture.

4. Based on your measurement results, comment on the relation between the gain of a horn

antenna and the operating frequency.

5. According to your measurement results, comment on the relation between the gain and size of

a horn antenna at a fixed frequency.

6. Compare the gain value measured for the horn antenna having dimensions a=17.6 cm,

b=15.4 cm with the gain value of the same antenna measured in Experiment-1. Comment on the

results.



V. References

[1] S. Silver, Microwave Antenna Theory and Design, Dover Publications, 1955.

[2] C. A. Balanis, Antenna Theory Analysis and Design, 3rd Edition, John Wiley & Sons, 2005.

Documents

EE426 LabMan Exp1 5 Spr2013