TW Lecture1

7/29/2019 TW Lecture1

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OFDM-MIMO implementation

in Line Of Sight

microwave/millimeter wave

link

Baruch Cyzs

[email protected]


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Introduction

Implementation of OFDM-MIMO in line of

sight microwave link

Description of hardware prototype of mmwave PTP microwave that employs

OFDM-MIMO.

Important Implementation issues inmicrowave link that employs OFSM-MIMO


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The MIMO Spatial multiplexing

implementation The MIMO spatial implementation exploits

random independent and identical distributed(iid) channel.

The orthogonality of the channel is usuallyachieved by existing of reach scattering.

Spatial multiplexing suffer degradation in itsperformance if significant direct path (LOS)

exists in the Rician channel. LOS microwave link cannot implement MIMO

since it relies mainly on strong LOS component


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How can MIMO implemented in

LOS Microwave For 3 decades LOS microwave links use polar

multiplexing by transmitting via orthogonalpolarizations.

Witcom in 2001 has initiated new activity ofimplementing geometric spatial multiplexingproject.

Prior to project kickoff Witcom has initiated

extensive outdoor field test to evaluate MIMOperformance in 5.8GHz in Tel Aviv.

Test results has shown low rank (mostlysingular) channel even in near/non line of sight.

The results has driven Witcom to seek solutionin the geometric spatial multiplexing.


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MIMO SM field test in 5.8GHz


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The antenna array approach

As opposed to polar multiplexing in spatialmultiplexing the number of SM channels

can be greater than 2. The LOS microwave link multiplexing

employs antenna arrays at both sides (noneed to be equal number of elements)

The array antenna spacing is the keyfactor for achieving orthogonality.


8/52

The Near Field multiplexing

The receiving array is located in the near

field of the transmitting array.

Since the wave front is not planar there isphase gradient upon the receiving array.

If the phase gradient is set to certain

predetermined value the link channelbecomes orthogonal.


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Geometry orthogonalization

R

R


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Linear antenna array requirement full

rank conditiond

R

R

dR

R

Phase difference between R and R:

360/(2*n) in optimal orthogonal condition

n antennas

1 4

2

opt

nRd

n

n

R


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The asymmetric case

dtR

R

R

R

n antennas

n

Rdd rt

dtdr

dr


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Optimal antenna spacing versus link

distance and frequency


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Singular values of dual array acts as

virtual channel gain

3 dB gain

optimal

Antenna spacing


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The optimal orthogonal case

characteristics Low sensitivity to antenna position.

No sensitivity to transversal shifts.

It is possible to work in suboptimal spacing

by employing adaptive modulation.

Antenna constellation can be linear or

regular polygon the same antenna spacing

rule holds.

( ) * ( )y t H x t( ) * ( )y t H x t * ( )y H x t


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h11

The channel Measurement

TX

TERMINAL

RX

TERMINAL

X1

X2

X3

y1

y2

y3

( ) ( )y t H x t( ) ( )y t H x t ( )y H x t

Measuring H matrix by a training/pilotsequence and calculating beam formers terms

for the channel separation


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Inherent diversity gain Apart of spatial multiplexing Beam formers exhibits

inherent diversity gain over SISO channel The gain depends on

Nt transmitters

Nr, receivers

Nc active sub-channels, for inherent systemgain:

10*[log( ) log( ) log( )]g Nt Nr Nc

NcNcNt

Nr

2 2


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Singular Value Decomposition

y

2

3

2

3

z

zUxVyU

zxVUyzHxy

domainf requncy

tztxthty

domaintime

HHH

H

)()(*)()(

X


18/52

The de-multiplexing process

3

2

1

3

2

1

3

2

1

3

2

1

'

'

'

'

'

'

'

'

'

z

z

z

x

x

x

y

y

y

Noise statistics has not changed (unitary rotation)

Singular values represent virtual gain


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Graphical presentation of SVD

Encoding

&

Modulation

V

+

+

Z1

Z4

Decoding

&

DemodulationU

V U

xx yy

Diversity

gain

Carrier

separation

1

2

Precoding is needed for diversity gain


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Basic Block diagram - dual

antenna arrays

V21

V12

V22

V11 U11

U21

U12

U22

x1

x2

y1

y2

H11

H22

H21

H12

x1

x2y2

y1

Tx Beam former

Diversity Gain

Rx Beam former

SeparationChannel


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Capacity discussion - theoryTheoretical capacity

2 options:Transmitter knows channel state:

1log( )

n

i

W

C

total

0

i

P

N

Where satisfies

1( )

i

i

Water filling algorithm

1

log(1 )n

i

Cn

i

Transmitter does not know channel state:


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Capacity discussion in real life

Real modem has maximum throughput

so there exists maximum bound of

throughput for higher SNR values. It transmitter knows the channel it can

set the throughput accordingly in the

modulator.

This is in fact real-life water filling.


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The dual mode QR-SVD weight

computation In order to decouple beam former update instances inboth sides of the link coefficients was set by dualalgorithm.

Precoding V coefficient in transmitter updated in slowmanner (sigma-beam sterring effect) on diversity gainby SVD calculation exploiting slow return channel.

Receiver beam former is calculated by QRdecomposition in fast manner update locally at the

receiver (delta-null steering effect):

zUz

QRQRU

QRHV

HR

T

11)(


24/52

The QR-SVD characteristics

If preceding V is set by SVD result R becomes

diagonal with singular values at its diagonal.

In the case of the the optimal orthogonal spacingR becomes diagonal and V is not needed.

Off diagonal elements energy of R (upper

triangle) proportional to the non unitary noise

enhancement.


25/52

The spatial multiplexing

implementation Witcom has built in the first half of the decade a

prototype system that utilizes the LOS mm wave

MIMO technology. The project was calledTeraWave.

This system was tested with successful results

for 9 months in France Telecom site.

Unfortunately due to marketing reasons theprogram has discontinued.


26/52

Terawave outline architecture


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TeraWave general specification Frequency: 23GHZ

Bandwidth 28MHz.

Capacity: STM-4 (622MBS).

4 parallel channels 155MBS each employpolar+spatial multiplexing.

Modulation: OFDM 46 subcarriers/symbol up to128QAM.

Full pilot symbol every 16 OFDM symbol. Coding: Turbo Product Code

Outline: Full digital IDU connected via fiber to dualODUS direct mounted to dish antenna.

DSP calculates SVD/QR coefficients in zero forcingfashion.


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TeraWave gallery


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Test site in France


30/52

The Spatial/polar system

Beam forming for multiple spatial channels separation

OFDM optimized modulation for spatial system

Smart mux for payload delivery over multiple spatial

channels

encoder

encoder

encoder

IFFT

IFFT

IFFT

Modulator

beam

formaerModulator

Modulator

OFDM framing

Demodulator

Demodulator

Demodulator FFT

FFT

FFTbeam

formaer

decoder

decoder

decoder

OFDM synchronizer

Demux

Mux

Data inData out

Spatial architecture

Transmitter Reciver


31/52

MACframer

M

u

x

FEC

fr

amer

ARQ

memory

management

return

channel

SDH

Ethernat

Payload

QoS

Beam

former

TPC IFFTUp

converterIF / RF

OFDM

framer

spatial

channels

TPC

TPC

adptive modulation control

code rate QAM

Up

converterIF / RF

clock LO reference

Up

converterIF / RF

Fiber

channels

IFFTOFDM

framer

IFFTOFDM

framer

from receiver

TeraWave transmission system

ODUsIDU


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MACframer

M

u

x

FEC

fr

amer

ARQ

memory

management

return

channel

SDH

Ethernat

Payload

Beam

former

TPC FFTDown

converterIF / RF

OFDM

framer

spatial

channels

TPC

TPC

adptive modulation control

code rate QAM

Down

converterIF / RF

clock LO reference

Down

converterIF / RF

Fiber

channels

channel

estimator

FFTOFDM

framer

FFTOFDM

framer

to transmit

TeraWave receiving system

ODUsIDU


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34/52

Resource AllocationConstant throughput mode in (TDM radio):

Assign modulation modes so as to maximize gain margin (dB

above minimum S/N required for reliable communication).

Channels with gain margin below a threshold are turned off if

throughput can be maintained with fewer channels.

Variable throughput mode (in packet radio):

When all channels have minimum gain margin, reduce

throughput in order to maintain gain margin.


35/52

Beamformer ImplementationSVD loop

[V] [U']

SVD

/QRcalculator

Actual channel[H(t)]

[U']

[V] []

Insertpilots

Extractpilots

Reference

PilotGenerator

X

Virtual Channel

DATADATA

Pilot

Generator

Return

Channel

CPE PhaseNormalization

x

Random phase

(t)

Estimatedvirtual channel

transferfunction


36/52

Implementation issues in mm wave

OFDM-MIMO Significant dynamic issues have been found due to thelarge aperture of the antenna array and interruptions closeto the antennas that caused too rapid changes indifferential channel.

Differential phase noise due to separate LOs in ODUs.

High common phase noise due to large PLL factor in23GHz.

Larger back off due of OFDM compared to SC.

128QAM require for low implementation degradation CINRof above 35dB.

PTP microwave require 99.999% availability that reflectBER=10-12 which permits less than 5 minutes outage peryear!.

Interpolation filters degrade symbol+CP periodity - a factthat increased noise in higher BB frequency. Remedy:

Over sample OFDM, better recover of symbol timingphase.


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To grant TeraWave signal processing

means to combat the known OFDM

drawbacks in order to eliminate most ofthe inferiorities compared to single carrier

system.

The challenge was met!

The TeraWave challenge


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MIMO-OFDM Phase noise error

discussion In SISO OFDM channel error :

Close to LO carrier common phase error.

Far from LO carrier Inter carrier Interference

Spatial OFDM error suffers from uncorrelated noise: Common phase error cause CPE error in each modem

that can be corrected by CPE compensation.

Differential Phase error cause uncorrected cross-talk

between sub channels that cannot be compensated byconventional CPE.

Channel Doppler causes mainly differential phase

error


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Correlated phase noise -

analysis Terawave has high OFDM symbol rate. Most of phasenoise is CPE type.

In Terawave there are no pilots in OFDM symbol. Fulltraining symbol is transmitted every 12-36 OFDM

symbols.

Channel model acquired after fading average of the pilotreference.

Channel model update rate 300Hz.

Safe acquisition and tracking for 128QAM requiresintegrated RMS phase error of less than 3.

Stringent requirement for MM wave receiver withsynthesizer with integrated RMS phase noise from300Hz of less than 3 in conventional phase recovery .


40/52

Solution decision directed

CPE Correcting common phase error after equalizing

among all the sub-carrier in the OFDM symbol after

slicing each sub-carrier.

Can be done in both forward (correcting actual data advantage over SC modem) and backward feed.

Prone to slice error due to AWGN, channel cross-

talk ICI phase noise, channel behavior and non

linearityprocessing gain depends on number ofcarriers.

In Terawave simulation showes 30KHz dual order

loop bandwidth. Practical allowed integrated RMS

phase noise from 3KHz - 3


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The spatial phase error noise -

calculation

1

1

1

1

'

, 1

' ( ) ( )

' ( )

' ( )

: ( )

'

tr jj

r t

r t

r t

r t

r t

H U e U V e V

if

H U I j U V I j V

H I j U U V V

H I j U U V V

Define E error matrix

H I jE

U U V V

1

1

1

1

'

, 1

' ( ) ( )

' ( )

' ( )

: ( )'

tr jj

r t

r t

r t

r t

r t

H U e U V e V

if

H U I j U V I j V

H I j U U V V

H I j U U V V

Define E error matrixH I jE

U U V V

1

1

1

1

'

, 1

' ( ) ( )

' ( )

' ( )

: ( )

'

tr jj

r t

r t

r t

r t

r t

H U e U V e V

if

H U I j U V I j V

H I j U U V V

H I j U U V V

Define E error matrix

H I jE

U U V V


42/52

The spatial phase error noise

2X2 case2 2

11 2 2 11 1 2 11 1 2

2 222 2 2 12 1 2 12 1 2

* *212 11 21 1 2 11 21 1 2

1

* *121 21 11 1 2 21 11 1 2

2

( ) ( ) (1 )( )

( ) ( ) (1 )( )

( ) ( )

( ) ( )

t r t t r r

t r t t r r

t t r r

t t r r

e v v

e v v

e v v u u

e v v u u

Common

CPE error

Diff. CPE error

Diff, x-talk

error


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Spatial phase error -

requirement Differential spatial phase noise causes CPE error andleakage from other spatial channel (noise like).

Without treatment algorithm the error budget force RMS

integrated phase error requirement of less than 0.7

degree tough solution in MM wave.

Alternatives:

To use common RF Lo (main contributor) for all ODUs.

Implication on deployment.

To use ultra quite separate RF LO, with basic high

frequency (low phase error multiplication).

To add to decision directed algorithm that mitigate

differential phase error mitigation the CPE leakage

compensation.


44/52

Spatial phase noise

(CPE+leakage) mitigation

xd1 CPE

calc.

Antenna 1FFT

Antenna 2

FFT

decision

ycpe1

ycpe2

y1

y2cpe2

xd2

+

+u*11

u*12

u*21

u*22 CPE

decisioncpe1

2 112 12

1 2

* *212 11 21 1 2 11 21 1 2 1

1

* *121 21 11 1 2 21 11 1 2 2

2

1 211 1 *

2 11 21

22 2

1

( )

( )

( ) ( )

( ) ( )

( )

(

CPE

CPE

CPE

t t r r

xd

t t r r

xd

xd new old xd

xd new old xd

y CPE y

x decision y

y x

e ex x

e v v u u

e v v u u

eu u

*

12

*

11 21

)e

u u

ycpe2

1x

2x


45/52

Differential Phase and amplitude

change due to wind


46/52

Solution to dynamics

Phase tracking loop in receivers according

to master transmitter to avoid differential

phase error. Differential amplitude correction between

DSP calculation.


47/52

Phase loop for high dynamics


48/52

The amplitude correction

for high dynamics

u12

h21

h12

h22

12

u22

u21

h11

+

+

+

+x1

x2

u11

x2

x1X

X

XX

X

X

XXX

X

X

11

XX

22

X

X

21

2

X

X

1

211112121212212121121111111 )()( xuhuhxuhuhx

212122122222211212112222122 )()( xuhuhxuhuhx

leakage

Leakage path

Leakage path

Gain path

Gain path


49/52

Phase noise and hit resilience

Phase

rotator

Phase error

measureFeedback

loop

control

dela

y

Feed

forward

loop

control

Phase

rotator

Phase

rotatordela

y

Phase

rotator

H

V

H error

V errorinput

-2

filter

Mediu

mRing

filter

M

UX

comparator

To phase

rotator

in ut


50/52

The constellation before and

after

After Correction Before Correction


51/52

PAPR reduction in MIMO

Each FEC Block is interleaved among 4

channels.

Novel approach of multiplying output withunitary matrix .

Rotation is selected according to minimum

peak to average. 2-3 dB gain in this approach.


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