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THE SILICON MICROSTRIP DETECTOR

THE SILICON MICROSTRIP DETECTOR

Proefschrift

ter verkrijging van de graad van doctor aan de Technische Universiteit Delft, op gezag van de Rector Magnificus, Prof.dr. J.M. Dirken, in het openbaar te verdedigen ten overstaan van een commissie door het College van Dekanen daartoe aangewezen op dinsdag 17 november 1987 te 16.00 uur

door

Warner Rudolph Theophile ten Kate

geboren te Leiden elektrotechnisch ingenieur

TR diss 1587

Dit proefschrift is goedgekeurd door de promotor Prof.dr.ir. S. Middelhoek

So eine Arbeit wird eigentlich nie fertig, man muss sie für fertig erkldren, wenn man nach Zeit und Umstdnden das Möglichste getan hat.

Johann Wolfgang von Goethe.

aan vader en moeder

CONTENTS

CHAPTER 1 INTRODUCTION 1

CHAPTER 2 OVERVIEW 7

2.0. INTRODUCTION 7 2.1. HISTORY 7 2.2. DETECTOR OPERATION 9 2.2.1. Radiant - electrical signal conversion 9 2.2.2. Signal processing 11 2.2.3. Noise 13 2.2.4. Radiation damage 16 2.3. DETECTOR TYPES 16 2.3.1. Application 17 2.3.2. Material 18 2.3.3. Fabrication process 19 2.3.4. Lay-out 20 2.4. CONCLUSIONS 23

CHAPTER 3 FABRICATION 25

3.0. INTRODUCTION 25 3.1. PART I - THEORY 25 3.1.0. Introduction 25 3.1.1. The leakage current 27 3.1.2. Carrier generation 33 3.1.3. Characterization 36 3.1.3.1. Temperature dependence 36 3.1.3.2. Lifetime measurement 38 3.2. PART II - EXPERIMENTAL 39 3.2.0. Introduction 39 3.2.1. /F-characteristic 41 3.2.2. Temperature 43 3.2.3. Lifetime 44 3.2.4. Radiation 47 3.2.5. Other processes 48 3.3. CONCLUSIONS 50

vm

CHAPTER 4 OPERATION 53

4.0. INTRODUCTION 53 4.0.1. POSITION PRECISION 53 4.1. PART I - THE PULSE RESPONSES ON A DETECTOR'S STRIPS 55 4.1.0. Introduction 55 4.1.1. Theory 56 4.1.1.1. The equations 56 4.1.1.2. Discussion 60 4.1.2. Results 62 4.1.2.1. The model 63 4.1.2.2. The pulses 65 4.2. PART II - THE INFLUENCE OF THE STRIP WIDTH 71 4.2.0. Introduction 71 4.2.1. Numerical model 73 4.2.2. Surface charge 76 4.2.3. Field plates 79 4.2.4. Bias dependence 81 4.2.5. Disturbance voltages on the strips 83 4.2.6. Experiments 87 4.3. CONCLUSIONS 97

CHAPTER 5 READOUT 101

5.0. INTRODUCTION 101 5.1. READOUT SCHEMES 101 5.1.0. Introduction 101 5.1.1. The passive readouts 102 5.1.2. The active readouts 106 5.2. CCD READOUT . . 108 5.2.0. Introduction 108 5.2.1. Injection time I l l 5.2.2. Experimental results 114

5.3. CONCLUSIONS 118

CHAPTER 6 CONCLUSIONS 121

REFERENCES 125

ix

SAMENVATTING - DE SILICIUM MICROSTRIPDETECTOR . . . . 137

ACKNOWLEDGEMENT 140

BIOGRAPHY 142

CHAPTER 1 INTRODUCTION

Semiconductor materials have found a widespread use in nuclear physics for radiation-detection purposes. At the beginning of this decade (1980), recognition of their usefulness in radiation detection for experiments in high-energy physics led to a renewed interest in this type of sensor and stimulated internationally the initiation of many research programs in this field. Position-sensitive detectors, in particular, received a great deal of attention, as current microelectronics fabrication techniques enable the production of detectors with a very high position resolution. In other fields of science (e.g. medicine and nuclear physics) as well, solid-state position detectors are attracting an increasing amount of interest and are finding more and more applications [1.1].

Fig.1.1. A typical event in a high-energy-physics experiment.

In high-energy physics an experiment can proceed as follows. Stable particles, such as protons, are accelerated in a cyclotron. When they have enough energy, they are directed towards a so-called target (see Fig. 1.1.), usually a slice of beryllium. The idea is for the protons to collide with the beryllium atoms. Upon collision, all kinds of reactions can occur, as long as they obey physical laws such as the conservation of energy, preservation of quantum number, etc. In high-energy physics this collision is referred to as

2

an event. Because of the high initial energy of the proton other, less stable particles can be generated at the event. By studying the events which have occurred, the theory describing the possible reactions can be verified, new particles (predicted by theory) can be discovered, or, last but not least, the properties of known particles can be studied.

The experiment as outlined above is called a fixed-target experiment (Fig. 1.1.). In another type, the so-called collider experiment [1.2], the initial particle (the proton above) is not directed towards a non-moving slice of material, but is allowed to collide with an identical particle (commonly its anti-particle) of the same energy moving in the opposite direction. In this way, the center of mass is not moving, and all of the energy is available for generating new particles.

Many other kinds of experiments are, of course, performed in h igh-energy physics as well (e.g. studying photon-photon interaction). What is relevant at the moment is to understand where the position detectors come in. This will be explained by returning to a fixed-target experiment, in which the properties of a resulting unstable particle have to be measured, see Fig. 1.1. Because the particle is unstable, it will decay to other particles before it can be measured. Consequently, its properties have to be deduced from those of the resulting secondary particles. The existence of the unstable particle can be proven by showing that its secondary particles originate from another point than all the other particles generated at the event (Fig. 1.1.). (In addition, the lifetime of the short-lived particle can be deduced from the distance between these points [1.3].) Such a point is called a vertex, and the detectors which measure that vertex are referred to collectively as the vertex detector.

A vertex detector measures the position of the resulting particles generated during an event at different places, enabling their tracks to be reconstructed. If this reconstruction is carried out sufficiently accurately, the vertex of the secondary particles can be distinguished from that of the primary (the intersections of the solid and dashed lines in Fig. 1.1., respectively), thereby proving the existence of the short-lived particle. For a whole class of particles nowadays under study, the required accuracy for the vertex determination can be achieved when the position detectors have a position resolution on the order of a few micrometers [1.2,1.3].

As silicon is a suitable material for the detection of radiation, it is a likely candidate for the fabrication of such position detectors. Only the well-known IC-fabrication techniques are used. The silicon microstrip detector was the very first semiconductor detector to be made in this way [1.4]. It constitutes the subject of this thesis. Other detectors made with IC-fabrication techniques include the CCD-based detector [1.5] and the silicon drift chamber [1.6].

When an ionizing particle traverses through silicon, it loses energy by creating electron-hole (eh) pairs and heat; on the average 3.6 eV will be lost for every eh pair created [1.7]. The particle is measured by collecting this charge. This can be achieved by making a reverse-biased diode. The detector

3

is made position sensitive by integrating an array of strip-like diodes (henceforth to be referred to as a strip detector), because only that strip through which the particle traverses will give an output signal.

Due to their high energy the particles in a high-energy-physics experiment move so fast that they barely interact with the silicon. However, due to this high speed the way they interact is nearly the same for all of them, despite their diversity in physical properties. Therefore, no distinction has to be made when discussing them in relation to silicon detectors. In that sense, these particles are usually referred to as minimum ionizing particles.

Minimum ionizing particles generate about 85 eh pairs per micrometer when traversing silicon [1.8]. This means that a depletion layer of a few hundred micrometers is needed to achieve a satisfactory signal-to-noise ratio (the preamplifier noise has to be accounted for as well). Collection by diffusion is not efficient and fast enough to contribute to the signal of the required detector. A detector is depleted by raising the reverse bias voltage. However, the breakdown phenomenon limits this voltage and consequently the maximum size of the depletion layer. The only way to achieve large depletion layers is to use high-purity silicon (HP-Si). To deplete a wafer (of about 400 nm thickness) completely the silicon has to have a (net) doping concentration on the order of 10 cm (or one dopant to every 50 billion silicon atoms!).

Such high-purity ("detector-grade") silicon is available nowadays [1.9]. Due to the low doping concentration, some device parameters have unusual dimensions. The Debye length, for example, is about 6 ym. Similarly, the depletion layer width at zero bias is about 27 ^m. As a consequence, the current behavior of diodes fabricated on this material will be greatly governed by the generation-recombination in the depletion layer, instead of by the usual diffusion outside this layer.

Likewise, there are some unusual aspects in the fabrication of microstrip detectors. Firstly, an entire wafer is covered with strip diodes in order to obtain one device. It is still possible, nevertheless, to achieve a reasonable yield, as the structure of the microstrip detector is relatively simple (except the strip length-to-width ratio of a few thousands) [1.10]. The malfunctioning of one or a few strips will moreover not prevent the device from being useful as a position detector. Secondly, the wafer is processed on both sides (masking is only needed for the strip side of the wafer; the processing on the rear covers the entire surface). Two junctions are needed for a low leakage current, which in turn is required for a low noise. A detector is a kind of wafer-thick pin-diode, except that the latter, with a higher doping level, but also with an intrinsic layer even smaller in size, can be modeled as being totally depleted at every bias voltage, with a (nearly) zero gradient of the electric field, whereas a detector clearly cannot. Finally, due to their large size, strip detectors require their own "packaging" strategy [1.1,1.11].

This thesis deals with the silicon microstrip detector as it is applied in a high-energy-physics experimental environment (measurement of the position

4

of minimum ionizing particles). Experimental devices have been made on silicon having a resistivity of about 2000-5000 Ccm (n-type). The strip pitch is on the order of 20 /mi. ("Pitch" refers to the repetition length of the strip pattern, i.e. the distance between the axes of two neighboring strips. Similarly, the interspacing between the strips is referred to as the "gap".) The total area of the devices was made much smaller than the wafer size. This was done to save material as well as to be able to use standard DIL-housings for packaging. Clearly, the same amount of knowledge can be gained from a device having a few strips which are also shorter in length. Only when performing measurements in a real high-energy-physics experimental environment (such as the many that have been carried out at CERN, Geneva) is a smaller area disadvantageous. Throughout the thesis, the detectors are assumed to be of the junction type. Surface barrier detectors, which are based on Schottky diodes, exhibit higher leakage currents and are made with a lower reproducibility (especially with large areas). They are therefore of less interest.

Briefly, the thesis is organized as follows. Chapter 2 presents an overview of semiconductor radiation detectors. This paves the way for a discussion of the fabrication (Chapter 3), the operation (Chapter 4) and the readout (Chapter 5) of the microstrip detector. In the thesis less attention will be directed towards such topics as radiation damage, yield and reliability, noise, packaging, and electronics.

Chapters 3 and 4 constitute the main chapters of this thesis. They are devoted to the two attractive characteristics of a semiconductor detector: its potential high energy resolution and its potential high position resolution.

Chapter 3 deals with the first aspect. It is divided into two parts. The first offers a theoretical description of the leakage current, as a high energy resolution requires a low leakage current. The second part presents the experimental results obtained on detectors (diodes) processed in different ways, showing the influence of the process on the detector's leakage current and thus on the energy resolution.

Chapter 4 takes up the second, positional, aspect of the microstrip detector. This chapter is likewise divided into two parts, which are preceded by a short discussion of the position precision of microstrip detectors. In the first part the output pulses on the detector's strips when generated charge carriers are collected by the detector are calculated. A new calculation method has been developed to obtain these pulses. The second part analyzes the influence of the strip width on the detector's performance. This is carried out theoretically by numerical calculation of the potential and electric field distributions in the neighborhood of the strips, and experimentally by measurements on fabricated test structures.

Chapter 5, treating the readout of a strip detector, begins by evaluating the existing readout methods. These are divided into two categories: the passive and the active readouts. Passive readouts are based upon resistor-capacitor networks, whereas active readouts include some active components. Subsequently, the use of JCCDs for readout will be considered in some more

5

detail. In particular, the transit time needed for the signal charge to flow from the detector strip into the CCD channel will be examined. Some experimental results are presented as well.

Chapter 6 closes the thesis by offering a short summary of the conclusions reached in the previous chapters.

CHAPTER 2 OVERVIEW

2.0. INTRODUCTION

In this chapter an overview of semiconductor radiation detectors is presented [2.1]. Compared to the gaseous one the solid-state detector has many advantages, the most important being the high energy resolution obtainable [2.2]. In a semiconductor detector about 3 eV is needed for the creation of one electron-hole pair, whereas a gaseous detector needs about 30 eV [2.2]. This means that the same radiation will generate much more charge in the semiconductor detector and consequently that the energy of this incident radiation can be estimated much better (see also Sec.2.2.3.).

Some other advantages of solid-state detectors are their linearity of response over a large range, their fast pulse rise time and their high density, which lowers the range of the radiation particles, thereby also enabling the more energetic ones to be stopped [2.2]. Finally, special configurations can possibly be made with semiconductor detectors by using the fabrication technologies known from microelectronics.

In the next section, Sec.2.1., the history of the semiconductor detector is briefly reviewed. In the subsequent section, Sec.2.2., some general considerations concerning the detector operation are given. It is divided into four subsections, which treat successively the interaction of semiconductor material with radiation, some signal processing, the noise aspects and the radiation damage. Sec.2.3. deals with the various kinds of detectors which can be made with a semiconductor. It is also divided into four subsections, in which the detector application, the detector material, the fabrication process and the lay-out are successively discussed.

2.1. HISTORY

The history of the semiconductor detector starts in 1951 when McKay observed that in a point-contact germanium diode measurable signals were produced by impinging a-particles [2.3]. At that time semiconductor materials had been investigated for particle detection [2.4], but the detection principle had been based upon modulation of the crystal conductivity by the radiation. The reverse-biased diode, having a lower noise level and a faster response time, is clearly a much better device.

At the end of that decade the semiconductor detector had become the center of interest. Different groups were doing research on this subject in a friendly spirit of competition, which is nicely described in [2.4]. The 7 t h

Annual. National Meeting (USA) in October 1960 was entirely devoted to "Solid-State Radiation Detectors", that being the "most timely theme" [2.5].

The very first detectors were based upon pn-junctions made by diffusion

8

[2.6], but Schottky diodes (Au-Si or Au-Ge) were soon being investigated [2.7]. The possibility of detecting radiation by these devices was first reported in 1955 [2.4]. Because these surface barrier detectors, as they are usually called, do not have high temperature steps incorporated in their fabrication process, lower leakage currents could be achieved on these devices. Moreover, the dead layer was much smaller.

At this stage, only a-particles could be detected and the research was headed in the direction of lower noise levels and larger depletion layers to enable the detection of /3- and minimum ionizing particles. The technologies springing from this research consist of Li-drifted silicon, high-puri ty germanium, surface passivation and guard ring structures. The development of the ion-implantation technique also enabled a new processing technique for detectors [2.8,2.9]. This technique combined the advantages of the former two (junction-type detectors - which means low leakage current, better reproducibility and yield - low temperature process and thin dead layers) and led in the end to the state-of-the-art as set by Kemmer [2.10].

Not only the energy of the impinging particle, but also its position can be measured by a semiconductor detector (and both with a high resolution). The first position-sensitive radiation detectors were based upon resistive charge division [2.11]. The charge division was done in the undepleted bulk. As the characteristics of this layer could not be controlled well, the resistive part was soon made by evaporating or implanting the division resistors [2.12].

Along with the development of these detectors gains were made in the theory behind their operation, and it became clear that with resistive charge division good position resolutions could only be achieved with high energetic radiation [2.12]. As a result the strip and checkerboard detectors were invented [2.13], which were able to measure both the energy and the position (in respectively one and two dimensions) of low energetic radiation. The disadvantage of these detectors is the necessity of having an amplifier chain for every strip; clearly, some multiplexing scheme had to be developed. Matrix addressing [2.14] and resistive charge division (by external resistors) [2.15] resulted.

At the beginning of this decade there was a revival of interest in the semiconductor radiation detector, due to the need in high-energy physics for detectors with a very high position accuracy. A resolution of a few micrometers was needed, and this led to the introduction of the microstrip detector [2.16]. Its much more complicated readout problem demanded and got more attention and more sophisticated solutions were under study [2.17,2.18,2.19,2.20] (see Chapter 5). Due to the more mature state of microelectronics, other types of detectors were also invented, namely the CCD detector [2.21] and the solid-state drift chamber [2.22]. Today's state-of-the-art is depicted well by [2.23].

9

2.2. DETECTOR OPERATION

In this section some general considerations concerning the operation of a semiconductor detector are given. Firstly, the way in which the material interacts with radiation is reviewed; secondly, the signal processing, mainly the preamplifier configuration, is discussed; thirdly, the various contributions to the noise in a whole detection system are considered; and finally, some remarks about radiation damage are made.

2.2.1. Radiant - electrical signal conversion

Roughly speaking, there are three sorts of radiation: charged particles, uncharged particles and photons. To be detected by a semiconductor detector they should liberate electron-hole (eh) pairs in the semiconductor, which, upon collection, will produce an electric output pulse, enabling further processing. In this generation process the particle will lose energy and will eventually be stopped in the detector. The amount of generated eh pairs is a measure of the energy deposit by the particle. In silicon each eh pair represents 3.6 eV, and in germanium 3.0 eV [2.24].

RADIATION ON SILICON

IONIZATION

hot electrons (few keV) &

lat t ice vibrations (heat)

IONIZATION

1 f ree electrons (1.12eV)

& heat

Ultimately, 3.6 eV used per generated electron-hole pair

Fig.2.1. The interaction between radiation and silicon in general.

The first sort of radiation, the charged particle, is stopped mainly by means of the Coulomb interaction (other effects can be elastical scattering, the Cerenkov effect and Bremsstrahlung [2.24]). At such an interaction, Fig.2.1., a bound electron is liberated and brought into the conduction band; the semiconductor is being ionized. (For this reason this sort of radiation is often referred to as ionizing radiation.) According to the laws of conservation of energy and momentum, the freed electron will have an energy of a few keV. However, it will lose this energy very quickly by creating other eh pairs. Because in the course of this whole process, which takes only a few

10

picoseconds [2.2], energy will also be lost in the form of optical phonons [2.2], the mean energy ultimately represented by each created eh pair will be larger than the band-gap width.

In the second sort of radiation the neutron is the main representative. Because the particle is uncharged it cannot be detected by electromagnetic interaction. Therefore, it first has to create a charged particle, which then can be detected. For this purpose a coating can be deposited on the detector, which interacts with neutrons in such a way that charged particles will be created [2.24]. For example, the coating may be a polyethylene layer; the neutron can knock out a proton from this layer. (Because the proton has the same mass, it is even possible that all the neutron's energy will be transferred to the proton.) Another solution is to let the neutron initiate some kind of nuclear reaction in which charged particles arise [2.24].

The last sort of radiation, the photon, is characterized by the fact that it reacts only once and disappears after that reaction. A photon is not slowed down as a charged particle is. There are three types of interaction, every type significant in a special range: the photoelectric effect, Compton scattering and pair production [2.2,2.24].

The first effect dominates at the lower energies, from visible light up to about 100 keV. Upon a reaction a hot electron is created, and the photon is lost. The electron, falling down to the conduction band edge, will behave as a charged particle by creating other eh pairs, thereby enabling the photon to be detected. (In the case of visible light only one eh pair (of the energy of the band-gap width!) is created per photon. Therefore only a continuous flow of light, resulting in an output current instead of an output pulse, can be measured, whereas at the higher energies each photon can be detected separately.)

From 100 keV up to 10 MeV the Compton scattering dominates [2.24]. In this type of reaction not only a hot electron, but also a new photon is generated. If this photon escapes out of the detector, too low an energy estimate will be made of the primary photon. As a result, instead of a single peak a broad spectrum ending at the photon's energy will be seen in the energy spectrum.

At the highest energies the production of electron-positron pairs will occur. The energy of the photon should be at least 1.022 MeV, the rest energy of such an electron-positron pair. The remaining part is converted into kinetic energy of the generated pair. After their generation, the electron and positron will be slowed down as charged particles. The positron will be annihilated along with an electron from the crystal, thereby generating two 511 keV photons. If one or both are not absorbed, too low an energy estimate will again be made. Now, instead of a broad spectrum, two extra peaks will be seen in the energy spectrum: the single and double escape peak [2.2].

The probability of an interaction depends strongly on the kind and energy of the impinging particle [2.24]. Heavy ions of low energy have a high such probability and are stopped in the surface region of the semiconductor crystal (cf. ion implantation), whereas elementary particles at

11

relativistic velocities hardly interact at all. (In fact, these minimum ionizing particles, as they are usually called, generate in silicon about 85 eh pairs per micrometer [2.25], which means an energy loss of only 300eV//mi and for a 300/im thick detector a generation of only 4 fC signal charge.) Photons of low energy are also more likely to interact with the semiconductor than those of higher energies [2.24]. The required detection volume therefore depends on the radiation to be detected.

2.2.2. Signal processing

By collecting the generated eh pairs the particle is detected. This can best be achieved by making a reverse-biased diode. To ensure fast and total charge collection the device has to be fully depleted (of course, this is not necessary for the detection of low-range particles). Besides the fact that collection by diffusion is not fast enough, partial depletion will not guarantee a complete charge collection, which can result in a wrong estimate of the deposited energy, or, due to the noise level, in particles remaining undetected. Moreover, the chance of trapping is larger in the undepleted part. This will also reduce the amount of collected charge, because, compared with the filtering time constant of the electronics, the trapped charge will not be collected in time. Typical collection times in fully depleted detectors are on the order of 10 ns [2.25].

BIAS

radiaton

^

~L

detector &

amplifier

pulse

PULSE HEIGHT

ANALYZER

(A.O.C.1

M.C.A.

MEMORY counts

t

ENERGY SPECTRUM

channel no(Enercjy)

J Fig.2.2. The method of measuring an energy spectrum.

The total amount of collected charge is a measure of the energy deposited by the particle. As a detector has a capacitive impedance, this charge can be measured by measuring the height of the detector's output pulse. This is done in a multi-channel analyzer (M.C.A.), see Fig.2.2.: The detector is irradiated with particles, and the heights of the subsequent output pulses are measured. The whole spectrum of possible pulse heights (the window) is divided into channels. Each pulse from the detector is counted by the channel corre­sponding to it. In this way an energy spectrum of the radiation is obtained.

12

The signal from a detector is too small for direct processing and some amplification is needed. There are three main types of preamplifiers, Fig.2.3.: the charge-sensitive, the voltage-sensitive and the current-sensitive one. They will be discussed below.

Q6(t)

a)

f CD

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Q6(ö e v *

A N y

1

R2

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f 'W

Q6(t)

e c)

n ^ v o u t : R , 6 ( t )

Fig.2.3. Basic amplifier configurations: a) charge-sensitive amplifier, b) voltage-sensitive amplifier and c) current-sensitive amplifier.

Usually, the preamplifier is of the charge-sensitive type (Fig.2.3a.). As a result of its large input capacitance all the generated charge will flow from the detector onto the feedback capacitance. The output voltage is therefore linear to the amount of charge generated, the feedback capacitance being the proportionality factor.

The disadvantage of this type of amplifier is its speed. Its rise time increases linearly with the detector's capacitance [2.26,2.27]. The voltage-sensitive amplifier (Fig.2.3b.) doesn't have this drawback. The rise time is independent of this capacitance [2.26]. However, as the output signal now depends on the capacitance of the detector, this capacitance should have a high stability. This demand is met in totally depleted detectors.

13

For very fast applications the current-sensitive amplifier (Fig.2.3c.) can be used [2.28]. Here, the pulse from the detector still has to be integrated, which in fact is done by the filtering system [2.28]. In the case of reading out a multi-electrode detector such as a microstrip detector, this type of amplifier is even preferred over the voltage-sensitive one, because, by virtue of its low input impedance, it will decouple the contiguous electrodes [2.26],

Because semiconductor detectors have no internal amplification the preamplifier should be a low-noise one, especially in those cases where the signal from the detector will be low (e.g. detection of minimum ionizing particles). First of all, the input transistor is responsible for the amplifier's noise, and therefore has to be of the low noise type. Normally, a junction FET is used for this purpose, but when short filtering times are used, such as in high-speed situations, a bipolar (microwave) transistor can be advantageous [2.27,2.29,2.30].

In the second place, the feedback resistor contributes to the noise. This favors the use of the charge-sensitive configuration, in which this resistance is very large or even left out altogether. However, in the case of the voltage-sensitive amplifier the feedback resistors can be chosen in such a way that their noise contributions are not of primary importance [2.27]. The noise contribution from the feedback resistor in the current-sensitive amplifier can also be greatly reduced by the use of short processing times [2.27]. It has been demonstrated that comparable results can be obtained with all three configurations [2.31].

2.2.3. Noise

The effect of the noise will be a reduction in the energy resolution, seen as a broadening of the peak in the energy spectrum (Fig.2.2.). The energy resolution is expressed by the FWHM-value (Full Width at Half Maximum). As the whole noise process has a stochastic nature, the individual noise contributions have to be added quadratically:

A£t2

ot = AEÏ + A£22 + A£3

2 + (2.1)

The contributions to the noise can be attributed to three main sources: noise in the radiation itself, noise from the detector and noise from the preamplifier. These will now be discussed successively.

The first main source of noise is attributed to the stochastic nature of the energy loss process of an impinged particle. Being a Poisson process, the resulting widening in the energy spectrum is proportional to the square root of the number N of generated electron-hole pairs, or relatively [2.2,2.24]:

**i = IT- * - # . (2.2a)

14

where E is the energy deposited by the particle and e the creation energy for one electron-hole pair (3.6 eV in the case of silicon). Eq.(2.2a) demonstrates why semiconductor detectors have a better energy resolution than gaseous detectors or scintillators: Because the creation energy e is smaller, more charge will be generated, resulting in a lower intrinsic widening.

In practice, the observed fluctuations appear to be less than predicted by Eq.(2.2a) and a correction factor F (the Fano factor [2.24]) has to be added:

AE, = Ijf-. (2.2b)

The reason for this smaller width of the peak is due to the fact that the total amount of energy loss is fixed; the particle is completely stopped in the detector [2.24]. For silicon F is on the order of 0.1 - 0.2 [2.32].

On the contrary, when the particle is not stopped in the detector, but traverses through it (e.g. minimum ionizing particles), the peak will be broadened. A tail will appear on the right side (the higher energies) of the energy distribution. This effect is known as the Landau distribution [2.33]. It is due to the fact that the particle has only a few interactions with the semiconductor. At each interaction an electron of a few keV will be created (Fig.2.1.); therefore a reasonable signal will result and the particle can be detected. However, fluctuations in the small number of these collisions will have a significant effect on the energy distribution, resulting in the Landau broadening [2.24]. When the detector becomes very small an even wider spectrum results [2.34].

Noise from the detector is the second main source contributing to the total noise. It is mainly due to current noise. In a usual pn-diode, where the current-voltage characteristics are governed by the diffusion process in the quasi-neutral regions, this noise is equated for the reverse bias case as [2.35]

ID = 2qIAf, (2.3)

where / is the value of the leakage current and the other symbols have their usual meaning.

However, in the case of a detector, the generation-recombination in the depletion layer is the main source of leakage current and noise (Sec.3.1.). It appears that in this case Eq.(2.3) can still be used, albeit with the introduction of a small correction factor [2.36]. Eq.(2.3) clearly demonstrates that a lower leakage current will result in a lower noise contribution from the detector.

Other contributions to the noise from the detector can be attributed to the series resistance (not depleted bulk or contact resistance), \/f noise, surface noise, and noise due to other leakage sources. All these are negligible when the detector is made under the customary circumstances.

The last main noise source is formed by the electronics. These are mainly

15

Fig.2.4. Detector-amplifier circuit with noise sources.

the bias resistor and the input stage of the preamplifier. When the bias resistance is high, its contribution to the (current) noise is negligible compared to the other noise sources, as they are depicted in Fig.2.4.

The noise from the amplifier is usually described by a series voltage and a parallel current noise source (Fig.2.4.). It stems mainly from the input transistor (its bias and gate or basis currents) and the feedback resistor, Sec.2.2.2., [2.29,2.37].

Usually, the effect of all of the noise sources is expressed by the ENC (Equivalent Noise Charge), i.e. the amount of charge which the detector should generate in order to give the same output signal. The effect of the series voltage noise source can be translated via the detector capacitance (differentiation) to a parallel current noise source and then all the noise contributions can be added [2.29]. In this way an optimum filtering time constant for a large signal-to-noise ratio can be estimated [2.29,2.37].

This procedure also shows that the detector contributes to the noise through its capacitance. A low capacitance is favorable, because it reduces the contribution from the amplifier's series voltage noise source. This constitutes, next to that of obtaining large signals (Chapter 1), a second reason for requesting large depletion layers (however, not in the case of microstrip detectors, as their interstrip capacitance is much larger than their junction capacitance, Sec.4.2.6.). A drawback of a large depletion layer is the accompanying increase in the leakage current (Sec.3.1.) and thus noise. Therefore, depending on the shaping time constant, the noise level of the preamplifier and the actual height of the detector leakage current, an optimum depletion layer can exist for a given set-up, where the noise will be the lowest (vice versa, given a fixed bias voltage, an optimum shaping time constant will be found for minimum noise) [2.38].

16

2.2.4. Radiation damage

As damage by radiation is inherent in the use of detectors, radiation detectors will always have a limited lifetime. Except in generating electron-hole pairs, the radiation will displace some atoms in the crystal [2.39]. The amount and kind of displacements (clusters, etc.) depend on the kind of radiation: whether or not it is charged, its mass, its energy, etc. [2.24]. The resulting crystal defects will create levels in the band gap [2.39,2.40], which means that the leakage current (noise) will increase (besides a change in the effective doping concentration). There will also be more trapping of generated eh pairs, which will result in pulse broadening or inefficient charge collection [2.41]. (Especially in the case of the silicon drift chamber, where the collection time of the generated charge carriers is on the order of microseconds, charge can be lost by the increased recombination velocity [2.42].) With respect to the microstrip detector there is another radiation effect which occurs when the detector is inhomogeneously irradiated (such as in fixed-target experiments): due to the local change in the effective doping concentration a transversal electric field will arise in the detector, which will consequently disturb the detector's position linearity [2.43].

Usually, the effect of radiation damage will be noticed after some critical dose has been exceeded. The level of this critical dose depends on the purity of the crystal and on the kind of radiation [2.39,2.40,2.41]. Commonly, a temperature treatment will anneal the radiation damage, allowing the detector to be used for an extra amount of time [2.40].

Whereas crystal damage is the effect of radiation in the bulk, in the oxide layers it will be charge buildup [2.39,2.42]. As a result, MOS circuitry is more sensitive to radiation damage than its equivalent made in junction technology (e.g. JFETs and JCCDs [2.42,2.44,2.45]). Charge buildup is due to the fact that those holes created by the ionizing radiation that escape initial recombination will stay trapped in the oxide layer, while their counter electrons will be swept out of it [2.39]. Consequently, the threshold voltage will change, ending in a malfunction of the circuitry. (Logically, these considerations concern only the electronics placed in the neighborhood of the detectors, especially in the case of a collider experiment, where the circuitry will be more subject to radiation than in the case of a fixed-target experiment [2.42].) As for the microstrip detector, there are also some effects which can result from the charge buildup in the oxide layer [2.42]: the surface leakage current can increase and the interstrip resistance can decrease, both of which affect the noise performance.

2.3. DETECTOR TYPES

All semiconductor detectors are based upon the principle of a reverse-biased diode; the impinging particle generates electron-hole pairs, which are collected by the electric field. However, within this base a large diversity of detectors is still possible. The semiconductor material, the geometry, the

17

lay-out and the fabrication process have to be chosen. These will depend on the application for which the detector is to be used. These considerations will be discussed in the following subsections.

2.3.1. Appl icat ion

First of all, it is important to know what kind of radiation has to be detected. As heavy ions and low energetic particles have a short range, the detector should have a thin dead layer. (The diode structure can even be designed horizontally, if necessary, to enable the radiation to enter the (sensitive) depletion layer directly.) Minimum ionizing particles and 7-photons have a long range and absorption length, respectively. The window thickness is less crucial, but a large collection volume is required for these particles.

Secondly, the purpose for which the detector will be used has to be determined. That can be for energy measurement, position measurement, timing and particle identification.

When the particle is completely stopped within the detector a very high resolution energy measurement can be performed, due to the low ionization energy of the electron-hole pairs, Sec.2.2.3. [2.24]. When the particle cannot be stopped by the detector, either a telescope or a so-called calorimeter construction can be made. In a telescope several detectors are stacked on top of each other, thereby effectively increasing the stopping volume. Moreover, the signals from the different detectors can be used for correlation, etc. (rejecting wrong data) [2.46]. In a calorimeter several detectors are likewise stacked, although in this case there is h igh-Z material (e.g. plumbum, uranium) in between them, thereby enlarging the stopping power of the whole system [2.47].

Fig.2.5. Two ways to make a detector position sensitive: a) by integrating an array of strip-like detectors; and b) by contacting the detector on two opposite sides, enabling resistive charge division.

A position-sensitive detector is made by integrating an array of detectors (strip and checkerboard detectors) or by contacting the detector on two opposite sides, thereby enabling resistive charge division [2.12], see Fig.2.5. Two-dimensional position measurement can be achieved by two one-dimensional position detectors in the case of particles which traverse the

18

detectors. In the other case, however, the detector should indicate both dimensions at once. This occurs, for example, in tomography (medicine), where the xy-position of ^-photons has to be measured. The first case occurs, for example, in high-energy physics, where the particles traverse the detectors. (It is even important to have small detectors there, in order to avoid scattering.)

One way to identify a particle is by measuring its energy loss over a short distance dE/dx together with its total energy E. The product of these two identifies the particle [2.24,2.48]. Another possibility, only useful for non-relativistic particles, is to measure the energy and the time of flight between two detectors [2.24]. The mass can be calculated from the (kinetic) energy and the velocity, measured by the time of flight.

Because semiconductor detectors have a fast response, they can also be used for timing purposes, one of these being the time-of-flight measurement. The triggering function is another clear example. In coincidence or anti-coincidence measurements the timing aspect is implicitly used [2.24].

2.3.2. Material

The most important materials for semiconductor detectors are silicon and germanium, as, due to the electronics industry, they are available in large quantities with a high purity and in monocrystalline form. As silicon can be oxidized, its surface is easy to passivate, thereby reducing surface leakage currents. Because silicon has a larger band gap than germanium, silicon detectors can be operated at room temperature, whereas germanium detectors will exhibit too large a current at such temperatures and will need cooling. A silicon detector has the extra advantage of having some electronics potentially integrated along with the detector [2.49,2.50].

Due to their higher density, solid-state detectors have a much better stopping power than gaseous detectors [2.4]. In this respect germanium is even preferable to silicon. Its higher atomic number (Z) implies a much better stopping power, as most of the stopping processes are proportional to a power of Z [2.24]. Therefore, research has also been done on other semiconductor materials with even higher Z [2.51,2.52,2.53]. The main problem is to fabricate these materials monocrystalline and with a high purity. For example, Hgl2 is a soft material, needing special care during cleaving and sawing to avoid mechanical damage [2.51].

Another material which has been studied is diamond [2.51,2.52]. Since this material is an insulator, it will have a low "leakage current", which means a low intrinsic noise level. However, its low Z, small size and high price prohibit wide-spread use. "Materials" with a very small band gap (and therefore possibly a high energy resolution) are found in the superconductors [2.54].

To achieve large depletion layers the net doping level of the bulk material should be as low as possible. This means that high purity or compensated material has to be used.. Compared with compensated material,

19

better diodes can be made on high-purity material, as this type of material contains fewer impurities, which means fewer generation-recombination centers and thus lower leakage currents (Sec.3.1.). Moreover, there is no chance of a redistribution of the doping (compensating) atoms, which would of course deteriorate the detector's performance. However, the fabrication of high-purity material is more sophisticated, and, due to the lower impurity level, it is also less radiation-resistant. Therefore, the purity grade should not exceed the needed level. Nowadays, nearly all desired levels of purity can be achieved. Only for really large depletion layers must lithium-compensated material be used.

High-puri ty silicon is fabricated by using starting material which already has a high purity grade. The final crystal is grown from these selected pieces. Then by zone refining the impurity content is diminished until the required purity grade is achieved [2.55]. As boron remains in the silicon during zone refining, p-type silicon will have a lower compensation level than n-type silicon of the same doping concentration [2.55]. The state-of-the-art is a doping concentration on the order of 1.10 cm (which means one dopant atom per cubic micrometer!). Lower concentrations are available, but difficult to achieve. As stated in [2.55], "business in this range is more like gambling."

Instead of using purifying techniques to achieve low doping levels, compensation techniques can also be used. In this way a net low doping level will result. The most successful method is based upon the lithium-drift process [2.2,2.24]. As an interstitial, lithium has a donor character. In the process it is brought into p-type silicon by a deposition step as in a normal diffusion process. In this way a pn-junction is formed. Then by reverse biasing and by heating, the lithium will drift to those places where boron atoms are left [2.24]. A perfect intrinsic region will result and a large depletion layer of several millimeters can be formed [2.55]. Due to the larger depletion layer (and higher impurity level) the leakage current will be higher, thereby increasing the detector noise. As a consequence, these detectors must be operated at low temperatures. Lithium-drifted germanium detectors even have the drawback of having to be maintained continuously at liquid nitrogen temperatures to prevent the. lithium from further diffusion and precipitation [2.2].

Another way, only suitable for silicon, is based upon neutron transmutation [2.24,2.56]. The (high-purity p-type) silicon is irradiated with neutrons. In this process some silicon atoms will get heavier because of absorbing these neutrons. When their atomic weight has grown to 31, they will disintegrate into a phosphorus atom by means of y9-decay [2.56]. In this way high resistivity uniformly doped n-type silicon can be fabricated.

2.3.3. Fabrication process

Based upon the different existing fabrication processes, it is possible to classify the detectors into three main types, Fig.2.6., namely the diffused

20

3|lm boron diffusion 30nm gold evaporation

n-Si n-Si

3|!m phosphorus

diffused detector

100nm aluminum

surface-barrier detector

.tLim boron implantation S3 £SS

n-Si

3u.m arsenic

ion-implanted detector

Fig.2.6. The three main types of detectors: a) the diffused detector, b) the surface-barrier detector and c) the ion-implanted detector.

detector, the surface-barrier detector and the ion-implanted detector. Diffusion was the first technique available for detector fabrication. The

high temperature steps necessary for the processing, however, degrade the material in such a way that the carrier lifetime is reduced, resulting in higher leakage currents (cf. Sec.3.2.). Moreover, due to the diffusion, the detector will have a reasonable dead layer.

Surface barrier detectors, on the contrary, are made in a low temperature process and can have extremely thin windows [2.57]. Instead of using a pn-junction, they are based upon the rectifying properties of the Schottky barrier. In practice, however, this metal-semiconductor interface is difficult to control, so that it is hard to make such detectors in a way easy to reproduce. Moreover, the (theoretical) leakage current of these devices is higher than that of the junction ones (made by implantation).

The ion-implantation technique offers the advantages of both techniques combined. Junction detectors can be made in a process with hardly any high temperature steps (cf. Sec.3.2.). Thin dead layers can also be made [2.10,2.58].

2.3.4. Lay-out

The most straightforwardly designed detector is the planar one, either square or circular in form. Using photolithography a guard ring can be designed around the detector area in order to lower the leakage current [2.59].

p

n-type

/'

bulk

))

/

Fig.2.7. The coaxial detector.

21

To achieve larger detection volumes, the detector can be made transversal. The best form in which this can be achieved is the coaxial detector, Fig.2.7. [2.60]. The surface in the hole forms one of the contacts, and the outer side of the crystal the other. In this way active volumes on the order of 100 cm3 can be achieved [2.60]. As the stopping power of these large volume detectors is of primary importance, these detectors are usually made of lithium-drifted or high-purity germanium.

A more refined lay-out is needed for position-sensitive detectors. The first detectors of this kind were based upon resistive charge division (Sec.5.1.1.), but, as it soon appeared, a better performance could be achieved by using array-type detectors [2.12], the checkerboard detector being the first of this kind [2.13].

strip detector drift chamber CCD detector

Fig.2.8. The three main types of high precision position detectors: a) the microstrip detector, b) the drift chamber and c) the CCD detector.

In high-energy physics detectors with a very high position resolution, on the order of a few micrometers, are needed (see Chapter 1) [2.41,2.42]. Up to the present three kinds of silicon detectors which have such a high position resolution have been invented, Fig.2.8.: the microstrip detector, the silicon drift chamber and the CCD detector. Another solution which also uses silicon consists of the scintillating fiber with silicon readout (e.g. photodiodes or CCD camera) [2.25,2.42]. As this detector is not a full-silicon device, it will not be discussed here. Additionally, based upon the drift chamber, a lot of other silicon detector types have been derived by implementing microelectronics technology [2.61].

The first of these to be invented, the so-called microstrip detector (Fig.2.8a.) [2.16], is made by the same process as the planar detectors, except that another masking set has to be used. The strip pitch of these devices is on the order of 20 /zm, resulting in a position resolution of a few micrometers [2.62].

Every strip has to be read out (see Chapter 5, which is the subject of this topic). This means that a lot of amplifier chains are required. Furthermore, the connection between the electronics and all those strips poses a technical problem. The easiest way to overcome these is to use charge division, either resistive or capacitive [2.12,2.25,2.42]. A more sophisticated approach is to use a second metalization layer, so that capacitive multiplexing can be performed [2.63]. The solutions based upon the full electronic multiplex

22

systems which have been invented so far are based on sample-and-hold principles [2.19,2.20] or on the use of CCDs [2.17,2.18,2.44]. These systems, however, do not eliminate the interconnection problem. Solutions to this problem are based on either the use of sophisticated bonding schemes [2.20,2.64] or on the integration of the electronics on the detector wafer [2.49,2.50].

A completely different approach for detectors with a high spatial resolution has resulted in the so-called drift chamber (Fig.2.8b.) [2.22]. In this case strips are made on both sides of the detector (which thus requires a double sided masking technique). Their doping is opposite that of the bulk material. There is only one strip (or point) having the same doping, which serves as bulk contact. To operate, the detector has to be fully depleted and a drift field towards the bulk contact has to be created. This is achieved by first reverse biasing the structure into full depletion and subsequently imposing different voltages on all the strips (increasing to the outer strips). It is also possible to bias only the outermost strips and let the others float [2.65]. (When, during the depletion process, the expanding depletion layers on both sides meet, the capacitance will decrease dramatically [2.22]. In this way, large sized, low-noise photodiodes can be made [2.22,2.25,2.66].) When electron-hole pairs are generated in the silicon, the ones will be collected by the strips directly, and the others will drift to the bulk contact. When the latter enter the neighborhood of that contact a pulse will be induced. The time difference between this pulse and the one from the strips indicates the drift time and therefore the position of the impinged particle [2.22]. With respect to the timing aspects fast, low noise electronics are required for a high performance of this type of detector. Moreover, the temperature should be stable, as the mobility of the charge carriers and thus the drift time depends on it.

Finally, the CCD detector (Fig.2.8c.) can be used for high-precision position measurement [2.21]. It is a two-dimensional position-sensitive detector which exploits the achievements of microelectronics. It functions in the same way as the CCD camera [2.41,2.67]: the electron-hole pairs generated by a particle are collected by a pixel (picture element) and the position is determined by successive readout. Due to its low noise, this type of detector doesn't need a large signal and therefore not a large depletion layer (this is still in reference to the detection of minimum ionizing particles, i.e. the particles in high-energy physics) [2.41,2.68]. However, cooling to liquid nitrogen temperatures is a prerequisite. Due to the development of fabricating and processing high-purity silicon for microstrip detectors, research has been started to investigate the properties of CCDs made on this type of silicon [2.45]. Because they have a large depletion layer, there can be no smear out of generated signal charge collected by diffusion, which would deteriorate the measurement. The output signal will clearly be larger.

In comparing the strip detector, the drift chamber and the CCD detector (Table 2.1., [2.25,2.41,2.42,2.69]), it can be concluded that the strip detector has the highest resolution and is the fastest device. However, the readout is

23

Table 2.1. Comparison of the strip detector, drift chamber and CCD detector.

properties

position precision (/jm) two-particle resolution (/im)

readout time

active area (cm2) fabrication technology notes

strip detector

2.5 60

(using cap. charge division)

10 ns (without multiplexing)

50 relatively simple one-dimensional

drift chamber

4.5 200

2 /is

50 complicated

needs uniform doping of bulk

CCD detector

4.3 40

40 ms

1 very complicated

needs cooling; thin active region

complicated and a lot of electronics are needed. The CCD detector, on the other hand, fits in well with the electronics and the interconnection. Besides, it provides two-dimensional position information. Drawbacks, however, are the long readout time, which doesn't allow high event rates (especially when the CCD remains active as a detector during readout), and the complicated technology, which doesn't allow the fabrication of large detection areas. When made in MOS technology the radiation resistance will also be lower [2.45] (Sec.2.2.4.). The drift chamber stands somewhere in the middle with respect to the other two devices (Table 2.1.). Which detector is going to be used depends strongly on the experiment (event rate, multiplicity of decays, etc.) for which it has to be used.

2.4. CONCLUSIONS

An overview of semiconductor radiation detectors has been presented. After a brief sketch of their history, the principles by which they can detect radiation were outlined. First, the interaction with radiation, which enables the semiconductor to be used for radiation detection, was summarized. Thereafter, the signal processing, mainly the preamplifier, was taken into consideration. And finally, the noise aspects of an entire detection system were discussed, followed by some remarks about radiation damage.

Next, the various types of semiconductor detectors came up for discussion. Even though the principle of operation is the same for all of them, there is still a large diversity among the detectors, depending on the application for which the detector has to be used. The crucial parameters which have to be selected in order to achieve that purpose are the semiconductor material, the geometry, the lay-out and the fabrication process. The large variation in these parameters leads directly to the large variety of semiconductor radiation detectors to be found in the literature today.

24

CHAPTER 3 FABRICATION

3.0. INTRODUCTION

Once the HP-Si is available, the manufacture of detectors appears to be straightforward. Only two junctions (a pn- and a hi-) have to be made, so that the processing sequence is relatively simple. Moreover, compared to VLSI-circuitry the mask requirements for microstrip detectors are seemingly less stringent; a micrometer variation in the strip's dimensions can easily be tolerated. On the other hand, however, a wafer-scale device has to be made with a reasonable yield, and, what is more, it appeared that the resulting detector quality (denoted by its leakage current) is strongly dependent on the processing sequence used.

There are many conceivable processing sequences resulting in the required junctions. Some of these protocols have been selected and used at the IC-Workshop of the Delft University of Technology for the fabrication of test diodes. As stated in Chapter 2, Sec.2.2.3., for a low noise contribution the detector should have a low leakage current. Therefore, the process resulting in diodes with the lowest currents should be selected. Extra processing steps, needed for, say, integrating some (multiplexing) electronics with the detector, however, may influence the leakage current and thus the choice of the proper process. It is therefore important to know their influence as well.

In this chapter the effect of the processing is discussed. As the leakage current is the most important quality indicator in a detector, the measurements are strongly related to this parameter. In order to interpret the measurement data, one needs a model describing the reverse bias behavior of the junction diode. Because the diodes discussed here are made on HP-Si some care has to be taken when simply adapting the usual formulas. Therefore, the theory of the leakage current as applied to these diodes has been reexamined.

The chapter is divided into two parts. Part i , Sec.3.1., reviews the theory of the leakage current and related subjects. In Part II, Sec.3.2., the measurement results obtained on differently processed diodes are presented and discussed. They will show the dependence of the leakage current on the choice of process.

3.1. PARTI - THEORY

3.1.0. Introduction

The quality of a detector is characterized by its leakage current. A low leakage current means a low noise and thus a good quality. Measurement of this parameter is therefore an inevitable tool in detector research and development. In this part the theoretical basis on which this and related

26

Table 3.1.1. List of symbols.

Q e k "i n,p N An, Ap

A» D L W W» Wgen

Wlh T

G ■ HL

^HL.norm T EK E,<0) EF £r Ei V J

subscripts:

elementary charge permittivity Boltzmann's constant intrinsic carrier concentration electron, hole concentration doping concentration deviation of the electron, hole concentration from its equilibrium value carrier mobility carrier diffusion constant diffusion length layer width depletion-layer width generation-layer width wafer thickness lifetime net generation rate effective surface generation velocity of the high-low junction normalized effective surface generation velocity absolute temperature activation energy band-gap width at 0 K (1.2 eV) Fermi energy level trap energy level intrinsic Fermi level bias voltage current density

I, II, m, IV indicating region I, II, III, IV of the diode (Fig.3.1.1.) depl QN n, p dep,indep

indicating region II (the depleted part) indicating region III (the undepleted, quasi-neutral part) indicating electrons, holes indicating depletion layer dependent, independent component

27

measurements are interpreted is outlined. First, in the next section, the theory of the leakage current is formulated and then investigated further in the subsequent section .with respect to the carrier generation. In the final section, the temperature dependence and lifetime measurement will be discussed.

Before going into more detail about the analytical description let us shortly analyze the leakage current qualitatively. In a reverse biased diode this current is due to [3.1,3.2]:

- thermal generation in the depletion layer, - thermal generation outside this layer (in all three directions) until

typically one diffusion length away, - surface generation, - injection from the (metal) contact.

Optical generation and (avalanche) multiplication are assumed not to occur. These four leakage sources can be reduced by:

- using high-purity material (a low number of impurities and thus of generation - recombination centers),

- contacting the bulk by a high-low junction (diminishing the effective diffusion length and avoiding injection from the contact),

- designing guard rings (automatically included in strip detectors; suppressing the contribution by surface generation and lateral diffusion),

- passivating the surface (usually by oxidation; reducing the surface states and thus the surface generation).

As the generation is a thermal process a further reduction can be achieved by cooling the device. Finally, a proper processing sequence has to be chosen, as will be shown in Sec.3.2.

In the following theoretical description ideal junctions, i.e. those without pinholes etc., are assumed. The analysis is kept 1-dimensional, as the geometry of most detectors is such that this restriction is allowed. Calculations of the depletion layer of a diode with a surrounding guard ring made on HP-Si showed that the electric field beneath the diode obeyed 1-dimensional theory, as long as the guard ring had a width of at least 300 /zm. However, measurements on 20 /im-pitch strip detectors showed that the leakage current through an arbitrary number of strips was already proportional to that number of strips when they were guarded by their nearest two neighboring strips (i.e. a total of four strips needed for guarding; see also Sec.4.2.6.).

3.1.1. The leakage current

To calculate the current in a fixed area, one has to solve the current and continuity equations, which are, for the sake of completeness, given respectively below [3.3] (See Table 3.1.1. for the meaning of the symbols):

28

n+

T in

IV

Fig.3.1.1. Subdivision of the diode structure into four regions, according to which the leakage current is calculated.

Jn = qnnnE + QDn-^,

Jp = qHpE - qDpjg,

dt " + q dx '

32 _ r L^s dt GP q dx

(3.1.1a)

(3.1.1b)

(3.1.2a)

(3.1.2b)

To derive an expression for the leakage current the diode structure is divided into four regions as shown in Fig.3.1.1. The outer two regions represent the high-doped regions, and the inner two the depleted and undepleted parts of the low-doped bulk (the abrupt depletion approximation [3.3,3.4] is assumed to be valid). The bulk is assumed to be of n-type behavior (in the case of p-type behavior the theory holds analogously). Therefore, the interface of regions I and II forms a pn-junction, while that of regions III and IV forms a high-low (hi-) junction. As explained by Gunn [3.5], the former junction may be made nearly impermeable to majority carriers, while remaining completely permeable to minority carriers, whereas the latter may be made the reverse (nearly impermeable to minority and permeable to majority carriers), and thus together they will lower the leakage current [3.6]. A reverse bias voltage which is applied will appear across the pn-junction. The potential drop across the hl-junction is negligible [3.5,3.6]; region II is the depleted (space-charge) part and region III the undepleted (quasi-neutral) part. WD represents the thickness of the depletion layer, which is proportional to the square root of the applied bias voltage (Eq.(3.1.12) below). Wth stands for the thickness of the wafer (about 450 /mi).

29

In each region there is thermal generation of electron-hole pairs, which, when collected at the contacts (either by diffusion or drift), contributes to the total leakage current. This current can therefore be written as the sum of the currents of each separate region:

J = JY + Ju + J1U + 7 I V . (3.1.3)

Region I is an undepleted (quasi-neutral) region. Its contribution to the leakage current is therefore determined by diffusion of the minority carriers in this region. When an infinite surface generation velocity, a constant doping profile and no degeneracy effects are assumed, the contribution of this region to the leakage current can be calculated as the standard solution of the current and continuity equations (the electron concentration at the edge of the depletion layer is taken to be zero, C = An/r, and L = v^Jr):

Jj = qül D ( 3 . L 4 ) N L tanh(H^L)

All coefficients are, of course, those of region I (for electrons). In most cases it will be sufficient to use Eq.(3.1.4) for an estimate of the

current contribution from region I. However, for very thin layers this equation will have too large a current, due to the assumption of an infinite surface generation velocity. In these cases the real value of this parameter has to be accounted for, or the equation might even be replaced by that of the Schottky barrier.

Region II is the depleted part of the diode. Usually, the generation rate is assumed to be the same over the entire layer [3.7]. This assumption can be used in a first approximation, but, as will be outlined in the next section, some caution must be exercised [3.1,3.8]. Still, when this assumption is applied, the current contribution of region II can be denoted as the standard (thermal) generation current [3.3,3.4,3.7]:

Jn = g^WD. (3.1.5) rdepl

Like region I, regions III and IV are undepleted. So, once again the diffusion of the minority carriers (which are now holes) determines the contribution of these regions to the leakage current. Because of the different doping levels it is convenient to model the contributions of these regions independently. At the interface the continuity of the carrier concentration and current have to be introduced as boundary conditions. The interface is chosen to lie just on the lowly doped side. When the current contribution of the third region is modeled the continuity condition is fulfilled by

30

introducing an effective surface generation velocity SJJL defined by [3.3,3.4]

J(x0 = qLpfxOS^, (3.1.6)

where X; denotes the interface on the lowly doped side. With this parameter region IV is modeled by (cf. Eq.(3.1.4)) [3.9]

>HL D N,

L tanh(H^L) WIV

III. (3.1.7)

Here W is the width of region IV and the coefficients are those for holes in region IV. Similar to region I, the surface generation velocity (at the metal contact) is taken to be infinite, as constant doping and no degeneracy are also assumed. Of course, these effects can be included in the model [3.10]; however, as will be shown below, the contributions of the first and the fourth region to the leakage current are negligible, and therefore no corrections for these effects have to be made.

When calculating the contribution to the leakage current of region III it is convenient to normalize the parameter S^.

SHL.norm = ^HL I 7 (3.1.8) Mil

Then the contribution of region III can be calculated by again solving the current and continuity equations (in fact, the contribution of region IV is included in this expression) [3.2,3.8]:

s i n h ( ^ ) + Wormcc*h(^)

COSh(^h_D) + S H L i n o r m Sinh(_Ü!__D)

where the coefficients are those for holes in region III and Ap(WD) is the deviation of the hole concentration at the interface of regions II and III from its equilibrium value, which increases with increasing bias voltage from 0 up to its maximum

*P(WD) = - £ l . (3.1.10) Nm

Note that for high values of ^HL.norm (infinite generation velocity) Eq.(3.1.9)

31

Table 3.1.2. Data [3.11] chosen to estimate the influence of regions I and IV on the leakage current.

Q =

"i = NY =

A = L, = W1 =

#111

A l l Av ^ in L IV

wlv

1.6 10" 1 9C 1.5 10 1 0cm- 3

10 1 8 cm- 3

2 10 cm /s 32 nm (T = 1 /is) 2 jum

1 10 cm i i n 2 0 " 3

1 10 cm = 12.5 cm /s = 2.0 cm2/s = 0.11 cm (r = 1 ms)

9 /zm (r = 0.4 /zs) = 2 /xm

-

Ji = 1.8 10"1 2A/cm2

W ™ = 8-9 10"7

reduces to an expression similar to Eq.(3.1.4). The contribution of region II is the most important one. Ju is on the order

of nanoamps per cm2 when rdep l is about 1 ms. It can now be shown that the contribution of region I is negligible so that high doping effects in this region do not have to be accounted for: As can be seen from the data in Table 3.1.2. (taken from [3.11]), the current from this region is on the order of picoamps per cm2.

The order of influence of the hl-junction can also be estimated with the aid of Table 3.1.2.: ^HL,norm *s about 10" , which is merely due to the large ratio of N1U versus Nw (Eq.(3.1.7)). Therefore, it is not necessary to account for high doping effects in this fourth region either.

Now the importance of the hl-junction can also be demonstrated. The n+-doping on the back side is not only necessary for a good ohmic contact between the metal and the semiconductor, but also for reducing the leakage current part due to diffusion [3.6,3.8]. Because of the low effective surface generation velocity 5 H L n o r m , the current contribution of region III will be lowered, as can be seen' from Eq.(3.1.9). The effect, known in the field of solar cells as the Back Surface Field [3.6,3.10], is more pronounced for

32

diffusion lengths which are large compared to the substrate thickness. (For small diffusion lengths the hl-junction is unimportant [3.8], and Eq.(3.1.9) results in the usual expression for the diffusion current [3.3,3.4].) Besides this reduction in the diffusion current, the hl-junction is also responsible for the suppression of any injection from the contact. These effects will improve with increasing doping level and thickness of the highly doped region [3.6].

In conclusion, the leakage current of a diode made on a high-ohmic substrate can be modeled as the sum of a generation and a diffusion current (the former being the most important) [3.2,3.8]:

J ~ ^depl + " QN

= l"1 WD + Tdepl

where [3.3,3.4]

WD = ^ ^ b i a s -

q ApfWW f" tanh(!!lh^D) ^ (3.1.11)

(3.1.12)

For the sake of completeness it is mentioned that according to this model the leakage current will saturate at full depletion and will stay at this value upon overdepletion (of course, multiplication and breakdown will occur at higher voltages [3.3,3.4,3.7]).

In practice, when measuring the leakage current, it is more convenient to split the leakage current into a depletion layer dependent (J^ep) anc* a

depletion layer independent (^ndep) component (having different dimensions) [3.2], instead of describing the leakage current as the sum of a generation and a diffusion current. These are defined by

J = Jdep.WD + 7 i n d e p . (3.1.13)

The diffusion current, decreasing with the applied bias voltage, contributes to both terms, whereas the generation current only contributes to the first one.

The contribution of the constant term can be described by the slope of the double logarithmic plot of the /K-curve, to be referred to henceforth as the «-factor. A value of one half (cf. Eq.(3.1.12)) indicates that the constant term is not contributing to the leakage current (i.e. the diffusion current as a whole is negligible), whereas values lower than one half do indicate such a contribution.

As outlined in the next section, generation in the depletion layer occurs in a smaller region than the width of the depletion layer itself, but

33

approaches it with increasing bias voltage. This effect will result in an «-factor larger than one half [3.1]. In practice, such values will also appear, because the junctions are not that ideal and effects like soft breakdown occur.

3.1.2. Carrier generation

When both junctions function well, the leakage current is only due to carrier generation in regions II and III. The generation rate is characterized by the lifetimes in these regions, defined by [3.12,3.13]

Tdepl = Tf". *"QN = -Q- (3.1.14)

The lifetime of the depleted region is referred to as the generation lifetime, while the other (mostly expressed via the minority carrier diffusion length) is called the recombination lifetime [3.12]. Because their physical origins are different, both lifetimes can be quite different in magnitude [3.2,3.12] and must therefore be distinguished from one another. Depending on their actual value, current-voltage characteristic Eq.(3.1.11) can be simplified further. When they are on the same order, the contribution made by the diffusion current is negligible and only the standard generation current will result. But, as will be shown below (Eqs.(3.1.16) and (3.1.17)), the lifetimes are not necessarily equal, and therefore the diffusion current cannot be neglected a priori [3.2,3.8]. Especially at low voltages disregarding this current must be approached with some caution, as its contribution will then be relatively large.

When talking about "the lifetime", one usually means the minority carrier lifetime in the (quasi-)neutral bulk (i.e. the recombination lifetime). This is the parameter specified by the manufacturer of the silicon. In nearly all practical cases (e.g. switching devices or solar cells) this lifetime is indeed the lifetime of interest. The depletion layer is so small that, in contrast to the detector case, its contributions can be neglected and only the behavior of the carriers in the bulk or base is of importance. (Yet another difference is that the recombination velocity of an excess of minority carriers is usually observed rather than the generation velocity due to a paucity of carriers -implicitly denoted by the use of the word lifetime!)

Three recombination mechanisms exist [3.3,3.4,3.12]: Shockley-Read-Hall (SRH), Radiative (band-to-band) and Auger recombination. Inversely, in the case of generation, these three processes can be recognized as well. However, under dark and low-field conditions the latter two will vanish, as there will be no photon absorption and no avalanche multiplication [3.12]. So, only the SRH-recombination-generation mechanism is of interest here. As usual, only one trap level (whose height is independent of its charge state) is assumed in the following, as, in effect, it is a good representative of all trap levels combined [3.3]. In reality, several trap levels are incorporated, none of

34

which is homogeneously distributed. Accounting for this would unnecessarily complicate the expressions.

The Shockley-Read-Hall generation-recombination is formulated by [3.3,3.4,3.7,3.13]

7Z:2 - pn G = —-. 4 7 r , (3.1.15a)

«! = « i e x p ( ^ i ) , Pl = W i e x p ( - ^ 5 ) , (3.1.15b)

n = >hexp(EF>n~Ei), p = « : e x p ( - ^ P l 5 ) , (3.1.15c)

where rp 0 and rn0 are constants characterizing the dominant trap (effects like the Poole-Frenkel effect, occurring at strong electric fields [3.14] are excluded). | ET-Ei | is the energy difference between the energy level of that particular trap and the intrinsic Fermi level; £ F and EF p are the quasi Fermi levels of the electrons and holes, respectively.

In the undepleted region n=N, p=?i?/N-Ap. So, by substituting this in Eq.(3.1.15) and comparing the result with Eq.(3.1.14), one arrives at [3.13]

'QN = r p 0 ( l + e x p & ^ L » ) ) + ^ ( e x ^ - ^ y ^ ) ) . (3.1.16) KI K.1

Usually, the second and third terms in Eq.(3.1.16) can be neglected, but in the case of HP-Si some care has to be taken in doing this [3.13]. At room temperature the Fermi level will be only 0.11 eV beyond the intrinsic level for a doping concentration of 1.10 cm" (Eq.(3.1.15c), [3.11]). So, the two terms can only be neglected for trap levels within ±0.11 eV from the intrinsic Fermi level. Outside this region either the second or the third term has to be accounted for.

In the depleted region both n and p are zero, and a completely different expression for the lifetime will be found [3.8,3.12]:

rdepl = r p 0 ( e x p f e 5 ) ) + r n 0 ( e x p ( - ^ ) ) . (3.1.17)

At the edges of this layer either n or p will increase, such that the generation rate, Eq.(3.1.15a), will decrease (The abrupt depletion approximation is still used). As a consequence, the region of carrier generation Wgen will be smaller than the depletion-layer width WD [3.1,3.8],

35

for which it should be substituted in Eq.(3.1.5) and in the first term of Eq.(3.1.11). Its borders are marked by (cf. Eq.(3.1.15a)) [3.8]

n = f— , (3.1.18a) pO

p = r p 0 \ V " 0 ^ . (3.1.18b)

For | ET-El | > | ^F.QN'^i I ' where EFQN is the Fermi energy level in the undepleted region, Eq.(3.1.18a) cannot be solved, as it would then require an electron concentration larger than the doping concentration. In that case, the border of the generation layer coincides with the depletion layer edge [3.8]. This is the case in which either the second or the third term in Eq.(3.1.16) cannot be neglected.

The width of the generation layer is proportional to a part of the depletion layer width [3.8]:

Wgen ~ (WD-WJ, (3.1.19)

where Wi is a constant whose magnitude depends on the height of the trap energy level. For a level at the intrinsic Fermi level Wx, reaching its maximum, is equal to the depletion layer width at zero bias, and consequently, Wgen will obey Eq.(3.1.19) at all bias voltages [3.8]. In the other case, Wgen will first increase rapidly upon application of a bias voltage, until it reaches the relation of Eq.(3.1.19) [3.8]. Accordingly, a steep increase in the current will be observed at the start of the /K-characteristic. Being similar in effect, it can only be distinguished from a possible contribution by the diffusion current by its temperature dependence [3.8].

As expressed by Eq.(3.1.19) the correction of WD by Wgen in Eq.(3.1.5) and in the first term of Eq.(3.1.11) is especially needed at low bias voltages, whose range can extend to about 20 V for a doping concentration of 10 cm" , depending on the actual value of Wj [3.1,3.8].

To summarize, the leakage current is due to generation in regions II and III; the generation rate is expressed by Eq.(3.1.15a). Region II is of thickness WD and region III of Wth-W-Q. In a part of region II, which is of thickness ^gen> both the electron and the hole concentration can be neglected (n,p=0) and Eq.(3.1.15a) reduces to

G = W i + W i • (3.1.20a)

36

All of the generated carriers are collected by drift. In the remaining part of region II (should this exist), the electron concentration is increased in such a way that it can no longer be neglected (Eq.(3.1.18a): n>ni,p1; there is still depletion: n<N, p=0), resulting in a decrease in the generation rate:

G = r p 0 / / r B 0 f t • (3-l-20b)

Consequently, the contribution of this part can be neglected. In region III the electron concentration is increased up to its equilibrium; there is no depletion (n=N; pn=nf-NAp). The generation rate is now expressed by

Ap (3.1.20c)

The carriers are collected by diffusion. By substituting Eq.(3.1.10) into Eq.(3.1.20c) and by comparing Eqs.(3.1.20a)-(3.1.20c), it can be seen how, depending on the heights of the trap and Fermi levels, the different regions contribute to the leakage current (their width also has to be taken into account).

3.1.3. Characterization

The leakage current of a detector can be characterized by its magnitude at a certain voltage (e.g. at full depletion). More information is gained by inspecting the whole /F-curve (e.g. measuring the «-factor, Sec.3.1.1.). Other characterization methods concern the determination of the activation energy or of the lifetime. These will be discussed in the next two subsections.

3.1.3.1. Temperature dependence

To estimate the way in which generation in the depletion layer and diffusion just outside that layer contribute to the leakage current, one usually measures the activation energy £A defined by

J = 7 0 e x p ( 2 ^ ) . (3.1.21) kT

In a first approximation, for a dominant generation current £A will equal roughly half the band gap width, while for dominating diffusion current the full width will result [3.3,3.4] (cf. Eq.(3.1.27)). Of course, when both mechanisms contribute an activation energy in between these extremes will result.

37

The factors which contribute to the activation energy are those which depend exponentially on temperature. It can be deduced from Eq.(3.1.11) that these factors are the intrinsic carrier concentration [3.3,3.4], the lifetime in the depletion region (Eq.(3.1.17), [3.12]) and the deviation of the hole concentration at the edge of the depletion layer from its equilibrium value:

«i ~ ^rt^r^ (3J-22)

'dep. ~ C O S h ( ^ ) , (3.1.23)

Ap(WD) = ü£. (3.1.24)

Eq.(3.1.24) is used instead of Eq.(3.1.10), because at high temperatures the low-doped silicon becomes intrinsic. The electron concentration n can be calculated as:

n = N + p, (3.1.25a)

np = V , (3.1.25b)

TV + VN2+4n-2 n = + " ' . (3.1.26)

2

From Eq.(3.1.26) it can be deduced that the electron concentration equals the doping concentration up to about n = §.\N. For A = 110 cm" this implies that the temperature has to be less than 50 °C [3.11], At higher temperatures the silicon starts to behave intrinsically. Consequently, Eq.(3.1.11) is invalidated, because it was derived under the assumption of an n-type behavior of the lowly doped region.

With the definition of Eq.(3.1.21) and with Eqs.(3.1.22)-(3.1.24) the activation energy of 7dep l and */QN (Eq.(3.1.11)) can be estimated as [3.7]

£A,dePi = 7 2 £ g W + | f i r - 5 l . (3.1.27a)

£A,QN = Eg(0). (3.1.27b)

In line with the remark on Eq.(3.1.16), the value of £A.,QN c a n De larger than stated here. Depending on the heights of the Fermi and trap levels, the

38

lifetime in the undepleted region can be exponentially dependent on the temperature, and can therefore influence the activation energy of 7QN.

By differentiating Eq.(3.1.21) with respect to 1/kT and by using 7=7depl+/QN an expression of the total activation energy can be derived:

E = " depl A.depl + QN ^A.QN (3.1.28) ^depl + ^QN

from which expressions for / d e p l and 7Q N can similarly be derived. The total activation energy is temperature dependent, because the ratio

between /dep l and J Q N changes with temperature, due to the different activation energies of these two components themselves. As depicted by Eq.(3.1.28), it will increase with increasing temperature. Likewise, it will decrease with increasing bias.

3.1.3.2. Lifetime measurement

As stated in Sec.3.1.2., the leakage current can be characterized by the lifetimes of the depleted and undepleted region. In practice, it is more convenient to distinguish a depletion layer dependent and a depletion layer independent current component (Eq.(3.1.13), Sec.3.1.1.), the former of which can be characterized by a lifetime [3.2]

rd e p = j p _ . (3.1.29) •'dep

Many methods have been invented to measure the lifetime [3.15,3.16,3.17]. One which is applicable to unprocessed materials is, for example, the photoconductive decay method [3.3,3.18]. However, it measures the minority-carrier lifetime. For measuring the generation lifetime a depleted region is required, which therefore necessitates at least a pn-junction or a MOS-capacitor.

Methods using a pn-junction are mostly based upon the switching characteristics of that junction, e.g. reverse-step recovery [3.3,3.19], open-circuit voltage decay [3.20] and short-circuit current decay [3.21], although also other methods have been invented [3.15,3.16,3.17]. All of these methods, however, are designed for pn-junctions with a small depletion layer, so that the diode current is dominated by the diffusion of the (injected) minority carriers in the (quasi-) neutral bulk (which may be affected by the recombination properties of the bulk contact). They are therefore measuring the minority-carrier lifetime.

Lifetime measurements based upon a MOS-capacitor usually make use of the relaxation to equilibrium of the capacitor when it is forced into a deep depletion by a voltage step applied to its gate. The Zerbst method [3.22] is a

39

well-known example of this technique. Heiman [3.23] used the same response, but proceeded in another way with the measurement results. A modification was made by applying a voltage ramp instead of a voltage step on the capacitor's gate, which kept the capacitor in a deep depletion [3.24,3.25]. This simplifies the interpretational effort involved.

When measuring on HP-Si some difficulties arise. The depletion capacitance will have a considerable bulk resistance in series, for which, depending on the actual measurement mode, the measurement results have to be corrected [3.26]. Furthermore, the generation contribution from the lateral directions as well as the capacitance change due to the lateral expansion of the depletion layer cannot be neglected [3.26], and a guard ring inevitably has to be used. Consequently, the use of a pn-junction for measuring the lifetime is preferred, as in MOS it is far more difficult to obtain the same (deep) depletion conditions under the diode as well as its guard ring. It is shown [3.2,3.8] that similar to the Zerbst plot obtained from the pulsed MOS-capacitor, a so-called IW-plot can be obtained from the biased junction diode, from which slope the lifetime can be determined (cf. Eqs.(3.1.13) and (3.1.29)). In an IW-plot the leakage current of the diode is plotted as a function of the depletion layer width. The former is obtained from the current-voltage (IV-) characteristic and the latter from the capacitance-voltage (CV-) characteristic, as the capacitance directly indicates the depletion layer width [3.3,3.4].

3.2. PART n - EXPERIMENTAL

3.2.0. Introduction

In this part, the measurement results obtained on the differently processed detectors will be presented [3.27]. Intuitively, one is inclined to say that the most important characteristic is the behavior as a detector, i.e. the interaction with radiation. However, the result of such a measurement is strongly dependent on the actual set-up (shielding, cabling, preamplifier, etc.). More knowledge is gained by measuring the dependence of the leakage current on the applied bias voltage. This is because, in the first place, a high energy resolution is usually guaranteed by a low magnitude of this current (Sec.2.2.3.), while in the second place, this measurement better characterizes the device in electrotechnical terms. Other measurements characterizing a fabricated diode are the determination of the temperature influence on the leakage current, the capacitance and series-resistance as a function of the applied bias, and, related to the leakage current, the lifetime of the charge carriers. Their theoretical background has been discussed in Sec.3.1.

All of the diodes have been made at the IC-Workshop of the Delft University of Technology [3.27]. Most of these were square, measuring 2.5 x 2.5 mm2, surrounded by a guard ring of 500 fun in width. When not otherwise stated the measurement results presented are based upon diodes of this size (where the guard ring is biased by the same potential as the diode).

40

The silicon used was n-type (phosphorus doped) high-purity silicon grown by the floating zone technique by Wacker-Chemitronic GmbH [3.28]. The resistivity of the material was specified as being about 4000 flcm (a variation of ca. 35% across the wafer has to be expected [3.28]) and the minority carrier lifetime as being above 500 fis. The wafers, 2 inches in diameter and 450//m thick, were of <111> orientation. The front side was polished, while the rear was etched "damage free". During processing, silicon wafers of the more usual resistivity of 8 Hem (n-type) were also added to the batch high-purity wafers, in order to check the processing or to verify/ compare the measurement results when necessary.

All of the processes used can be characterized by four main processes, which are listed in Table 3.2.1.; all of the others are variations or combinations of these. Only the most important processing data are given. (For a more detailed description see e.g. [3.3,3.4,3.29,3.30,3.31].)

Table 3.2.1. The four main processes.

process

type

p+-layer

dopant

junction depth

concentration

n+-layer

dopant

junction depth

concentration

cooling sequence after last high-temperature treatment

annealing

I II

diffusion-based

standard

boron

. 3.0 fim , . 1 8 - 3 10 cm

standard

phosphorus

2.5 /im , . 2 0 - 3 10 cm from 1000 °C to 900 °C in 15 min. followed by another 15 min. staying at 900 °C

deep diffused

phosphorus

15 jim , . 2 0 -S 10 cm from 1000 °C to 800 °C in 1 °C/30 sec

III IV

imp Ian tat ion - based

5 1014cm"2, 40keV

boron

0.4 /im

i n 1 9 " 3

10 cm 5 1015cm"2, 150 keV

arsenic

0.3 fim , . 2 0 - 3 10 cm

600 °C 30 min. in a nitrogen ambient

900 °C 30 min. in a nitrogen ambient

41

Processes I and n are diffusion-based. Process I is based upon the standard diffusion process of the IC-Workshop as it is used for the fabrication of epitaxial npn-transistors (p+-diffusion for the base; n+-diffusion for the emitter). In process II the high-low junction on the rear side of the wafer is made deeper into the crystal. This has been designed with the idea of "shielding" the influence of this relatively rough surface. In all likelihood, the gettering will also be improved by this deeper diffusion. In this process the cooling sequence after the last high-temperature treatment is done much more gradually in comparison with the standard sequence of process I. This has been designed with the idea of enabling the crystal to recrystallize easily from its high-temperature disorder [3.17].

Processes III and IV are ion implantation-based. In these processes high-temperature treatments are avoided as much as possible, as it is believed that these will deteriorate the final diode characteristics [3.17,3.32,3.33,3.34], (see also Sec.3.2.5.). Only for the oxidation were high temperatures used, as the oxide layers were grown thermally. These processes were inspired by that of Kemmer [3.30,3.32]. The implant doses are chosen equal to Kemmer's and have not been varied in the experiments. So, the dose of boron has been beneath the critical dose and that of arsenic above it [3.29]. To reduce channeling effects the implantation was done through an oxide layer of about 0.04 urn in thickness. The implantation energy is chosen larger than Kemmer's. This was done to avoid possible problems with alloying (with respect to the arsenic this was also done to have the high-low junction, once again, as deep as possible). In the experiments, the annealing temperature was varied between 600 °C (this was inspired by Kemmer [3.32], but is also the lower threshold for low currents [3.29,3.35,3.36]), being process III, and 900 C (the usual annealing temperature for IC-fabrication [3.29]), being process IV.

In the next sections the measurement results will be presented and discussed. When not explicitly denoted, the data presented will always be taken from the same samples, in order to enhance the correlation between the different graphs. The samples are chosen so as to give a good picture of what was measured on the average and what is reproducible. Furthermore, they were chosen out of the same batch, which means that identical processing steps were carried out at the same time under identical circumstances.

3.2.1. /K-characteristic

Besides processes I and II, it is feasible to interchange the cooling sequence after the last high-temperature treatment, resulting in two new processes, which are labeled process I' (standard diffusion with slow cooling rate) and process IF (deep diffusion with standard cooling sequence). Figs.3.2.1a.-d. illustrate the /^-characteristics of diodes made by these four processes. The curves presented here are not the best which were achieved, but are instead good representatives of the average results (e.g. the best result achieved in process II is about 8 nA at 100 V). The measurements were

42

****** GRAPHICS PLOT ****** MA 3 5 1 5A

* * * * * * GRAPHICS PLOT * * * * * * HA 3 5 1 6 B

1

/ / /

/ /

c /

y /

S

a) VB 1 0 . 0 0 / d l v ( V)

10.00 /div

.0000

y

b) VB 1 0 . 0 0 / d l v | V)

x x x x x x GRAPHICS PLOT x x x x x x MA 3 5 1 2A

x x x x x x GRAPHICS PLOT x x x x x x MA 3 5 1 3A

10.00 /div

.0000 / /

/ f

/

c) . 0 0 0 0 1 0 0 . 0

va 10.00/dlv ( v)

a. 500 /div

.0000 / /

/ / / "*

d)

x x x x x x GRAPHICS PLOT x x x x x x IS 445 5

x x x x x x GRAPHICS PLOT x x x x x x IS 445 Z

1.000 /div

.0000

/ ,/

/ ' /

e) VB 1 0 . 0 0 , ' d l v ( V)

20.00 /Olv

.0000 / f

/ /

/

/ / s

y y

f) VB 1 0 . 0 0 / d l v [ V)

Fig.3.2.1. IV-characteristic of diodes made by a) process I (MA35I-5A), b) process I' (MA35I-6B), c) process II' (MA35I-2A), d) process II (MA351-3A), e) process III (IS445-5) and f) process IV (IS445-2). (Note the different scales.)

43

carried out at room temperature (under dark conditions). The results clearly demonstrate the superiority of process II over process I. Processes I' and i r give results in between these two. On the whole, it is believed that the slow cooling treatment is the most important action for lowering the leakage current.

Figs.3.2.1e. and 3.2.If. present the /^-characteristics of diodes made by processes III and IV. Once again, they do not represent the best results achieved, but rather the reproducible average (e.g. the best results obtained in process III were saturated currents of about 1 nA). The results clearly demonstrate the superiority of process III, not only with respect to the implantation-based processes, but also with respect to all processes (Bear in mind, as discussed in Sec.2.3.3., that implanted detectors are also favored for their smaller window).

3.2.2. Temperature

The leakage current dependence on the temperature was measured at different bias voltages. The measurements were performed in intervals of 5 °C from room temperature to the maximally allowed 50 °C (Sec.3.1.3.1.). The applied bias voltage was 10, 20, 40 and 80 V. The activation energy (Eq.(3.1.21)) at the different bias voltages was determined from the measured currents, along with the «-factor (Sec.3.1.1.) at the different temperatures. The results are depicted in Tables 3.2.2. and 3.2.3. (For process III the activation energy at low bias voltages (see below) is also presented).

Table 3.2.2. The measured activation energy (in eV).

process I

process II

process III

process IV

process HI

10 V 20 V 40 V 80 V

0.72 0.70 0.68 0.68

0.81 0.77 0.76 0.76

0.83 0.79 0.74 0.72

0.76 0.76 0.75 0.73

0V I V 2 V 4 V

1.10 0.97 0.94 0.88

It can be seen from Table 3.2.2. that the activation energy decreases with increasing bias, because, as explained in Sec.3.1.3.1., the ratio of ^depi/^QN increases. Similarly, the «-factor decreases with increasing temperature (J indep increases with temperature more than / d e p ) . It can also be seen that the

44

Table 3.2.3. The measured n-factor. (RT = room temperature.)

process I

process II

process in

process IV

RT

0.62

0.58

0.51

0.78

30 °C

0.58

0.56

0.51

0.76

35 °C

0.54

0.55

0.49

0.77

40 °C

0.50

0.49

0.42

0.72

45 °C

0.50

0.49

0.39

0.68

50 °C

0.46

0.48

0.38

0.68

low-current devices (those made in processes II or III) have a higher activation energy (at a small bias), indicating a relatively larger contribution from the depletion layer independent current (Sec.3.1.1.).

As the best diodes are clearly made by process III, their characteristics have been analyzed a little bit further. Out of all the diodes made, their /^-characteristic best fits the theory presented in Sec.3.1. (In the other processes the junctions are probably less ideal, and higher currents result). As can be seen from the plots, the leakage current first strongly increases when a bias voltage is applied. Thereafter, the square-root relationship, Eq.(3.1.12), holds quite well (the «-factor is about one half). The step is due to the diffusion component 7QN, Eq.(3.1.11), which increases very fast with the applied bias, as Ap(WD) increases from 0 to its maximum value expressed by Eq.(3.1.10). As explained in Sec.3.1.2., it could also be due to a steep increase in the width of the generation layer Wgen. Measurement of the activation energy at low bias voltages (see Table 3.2.2.), however, clearly showed the domination of the diffusion component. (The activation energy at 0 V is determined from the temperature dependence of the depletion layer independent current component, Sec.3.1.1., which in turn is determined from the /(^-characteristic, which is presented below, Sec.3.2.3.) Assuming that at large bias voltages the leakage current only consists of generation current, it can be deduced from Table 3.2.2. that the dominant trap level is about 0.1 eV from the intrinsic level (Eq.(3.1.27a)).

3.2.3. Lifetime

The /^-characteristics were also analyzed by determining the lifetime [3.26], defined by Eq.(3.1.29), and which, as explained in Sec.3.1.3.2., is determined from the slope of the IW-characteristic. For devices made by processes I-IV such characteristics are shown in Fig.3.2.2. These devices were made in a different batch, so that not all of the data given above applies. (All of the data presented in this section are from this other batch.) The specification of the silicon used is the same as mentioned in Sec.3.2.0. Only its orientation, being <100>, and its resistivity, which was specified to be about

45

c) Fig.3.2. process (IS464

2. VN -characteristic of diodes made by a) process I (IS464-4), b) II (IS464-1), c) process III (IS464-8) and d) process IV -6).

1600 ficm (still with a variation of about 35% across the wafer) are different. To obtain the /^-characteristic the CK-characteristic has to be measured

(Sec.3.1.3.2.). During such capacitance measurements the effect of the series bulk resistance has to be taken into account [3.26]. In Fig.3.2.3. this resistance is depicted as a function of the depletion layer width. As to be expected, it is decreasing with increasing depletion layer width; its extrapolated crossing with the horizontal axis is around the wafer thickness. A bulk resistivity could be deduced from the slope of this curve, which is depicted in Table 3.2.4. for the specific devices together with the value obtained from the slope of the C K-characteristic [3.3,3.4]. This latter curve was only a straight line when the guard ring was used for the measurement, which emphasizes its necessity. The data show that the resistivity of the wafers was rather around 2000 ficm, which is still within the specified toleration range. No processing influence on the resistivity can be deduced.

Table 3.2.5. lists the deduced lifetimes. The /^-characteristic of the samples from this batch showed a bend around 70-80 /jm, after which the

46

l.ZO

1.05

0.90

0.75

0.60

0A5

0.30

0.15

0.00 O

a) 1.B0

! 160

i t o

1.20

1.00

0.80

0.60

0.10

020

000

320 100

Wlllml '

1.50

1.20

0.90

0 60

0 . 3 0 -

0 00

b) 320

Wlllml -

c) d) Fig.3.2.3. The series resistance as a function of the depletion layer width. a) process I, b) process II, c) process III and d) process IV.

Table 3.2.4. Bulk resistivity (in kUcm) determined from a) depletion layer capacitance (C) and b) series bulk resistance (R).

process I

process II

process III

process IV

C

1.91

2.46

2.06

2.11

R

2.18

2.34

1.98

2.11

47

Table 3.2.5. Lifetimes (in us) determined from the IW-characteristics.

process I

process II

process III

process IV

Ti

493

636

8025

108

T2

36

346

2900

39

current increased more rapidly. Consequently, there are two lifetimes, TX and r2, characterizing respectively the first and the second part of the /W-curve. The reason for this bend is not well understood. It could be due to a kind of uncompleted gettering, although this is not likely, because the bend occurs at nearly the same spot, independently of the process used. Presumably, the difference was already present in the unprocessed wafers (as a kind of 75um-wide denuded zone). Like the /K-characteristics (Sec.3.2.1.) both lifetimes clearly demonstrate the superiority of process III, not only with respect to the implantation-based processes, but also with respect to all processes.

By substituting the trap energy level found in Sec.3.2.2. for the diodes made in processIII (0.1 eV = 4kT) in Eqs.(3.1.16) and (3.1.17) (the Fermi energy level is also about 0.1 eV beyond the intrinsic level for a doping concentration on the order of 10 cm , Sec.3.1.2.) and by comparing the results (assuming rn0=r 0), a ratio of about 25 will be found for r d e P I A Q N . This points in the same direction as the ratio of the measured value, Table 3.2.5., and the specification of the manufacturer.

3.2.4. Radiation

The detectors were tested with several kinds of ionizing radiation (cf. Fig.2.2.). As mentioned, such tests provide more information about the measurement set-up than about the detector. In Fig.3.2.4. the spectra obtained with 7-radiation from a Am-source (60 keV) are presented as an example. The same measurement set-up has been used throughout. The diodes, biased at 60 V, were all irradiated the same length of time (5 min.). A 0.1 mm-thick copper plate was placed in between the detector and the Am-source in order to suppress the X-rays at 20 keV. The shaping time was 1 us. The FWHM-values are depicted in the figure. The relation between the magnitude of the leakage current (Fig.3.2.1.) and the energy resolution is clearly demonstrated by this figure. (As a matter of fact, without the copper plate the 20 keV X-rays could only be detected with the detectors made in processes II or III, the best of course with the latter.)

48

11

10

9

e

7

6

5

2

$ 1

1 ■

-

MA 351 SA FWHM =S.3keV

1 A vy \ a)

IS « 5 5 FWHM = U.U keV

c) 241 Fig.3.2.4. Spectra from "Am (60keV ^-radiation), a) process I

(MA35I-5A), b) process II (MA351-3A), c) process III (IS445-5) and d) process IV (IS445-2).

3.2.5. Other processes

Besides the four processes listed in Table 3.2.1., a number of other processes have been tested [3.27], namely those which are more or less combinations or variations of the four main processes. Interchanging the cooling sequences in processes I and II has already been mentioned (Sec.3.2.1.). Similarly, with respect to processes III and IV annealing temperatures of 700 C and 800 °C have been experimented with. The resulting characteristics were in between those of the 600 °C and 900 °C devices. The difference between the 600 °C or 700 °C anneal was small (although the 600 C one still had the best characteristics).

Of more interest are the characteristics resulting when extra processing steps or a combination of steps are used. Such steps can be necessary when, for example, some (multiplexing) electronics have to be integrated on the same wafer as the detector (Chapter 5). In particular, there is a lot of uncertainty about the behavior of HP-Si during high-temperature treatments. The

49

resistivity of the silicon may change [3.37,3.38,3.39] or the final diode characteristic may deteriorate [3.17,3.32,3.33,3.34]. With respect to the resistivity, however, the measurements showed no such influence within the specified tolerance of 35% (cf. Sec.3.2.3., Table 3.2.4.), in contrast to the remarkable process influence on the leakage current.

«««Kult GRAPHICS PLOT x x x x x x IS 463 6

xxxxxx GRAPHICS PLOT xxxxxx IS 463 3

1.000 /div

.0000

1 J 1.000 /dlv

.0000 /

/

a) b)

10 «ui

-200.0

xxxxxx GRAPHICS PLOT xxxxxx IS 463 9

/ /

/ /

/ V •'

• y

s s

/

<• /

'

c) io.oo/mv ( v!

Fig.3.2.5. IV-characteristic of diodes made by a) process III (IS463-6), when b) dummy temperature treatments are applied beforehand (IS463-3), and when c) a high temperature (annealing) is applied afterwards (IS463-9).

First, dummy steps were applied on the wafers of one batch, which were processed according to process III, i.e. one wafer was processed (as a monitor) entirely according to this process, while the others, being oxidized, were first treated, in the furnaces as if a p - or n-diffusion or both was to be made. The resulting leakage currents were of the same magnitude; the curvature of the /F-characteristic was a little smoother for the dummy devices, see Fig.3.2.5. (The dummy treatment of the characteristic shown consisted of the extra oxidation followed by the necessary treatments for both p + - and

50

n+-diffusion formation from process I.) However, when a high-temperature step (e.g. annealing at 900 °C) was applied after the implantation, the /F-characteristic was severely deteriorated and resembled that of a diode made in process IV, whether or not a 600 °C annealing step was also applied (either before or after the 900 °C anneal). (Fig.3.2.5. shows the case in which a 900 °C annealing has been applied after the 600 °C annealing.) Consequently, when extra processing is required, the high-temperature steps have to be executed before the detector junctions are implanted.

After the 600 °C treatment the implantation damage had been completely annealed out [3.34]. This effect is also reported with respect to the implantation of boron at high energies [3.36,3.40]: After the 600 °C anneal the lattice damage is completely restored, although not all of the doping atoms are activated, whereas after the 900 °C treatment they are all activated, but the lattice damage reoccurs, this time irretrievably. In all likelihood [3.40], all of the boron atoms are then (at total activation) in substitutional lattice sites, thereby also increasing the lattice strain, due to the relatively large lattice dilatation. Before their activation, the boron atoms were located in interstitial lattice sites and, consequently, gave no (or less) rise to the lattice strain. As a result, along with the changeover of the boron atoms from the interstitial (inactive) to the substitutional (active) sites, dislocations are created, which are responsible for the increase in the leakage current. This also explains why an accompanying anneal at 600 °C after a 900 °C treatment has no effect on the final current (it will not cause the boron atoms to move from their substitutional sites back to the interstitial ones).

Finally, the effect of forming the junction by both diffusion and implantation has been studied. The protocols of processes I and III have been used. The wafers were all made in the same batch. There are nine possibilities, as each junction (pn- or hi-) can be made in either process or in both together. The resulting /^-characteristics showed, Fig.3.2.6., that the implantation process yielded much better diodes than the diffusion process alone (as already known from Sec.3.2.].), while their combined use resulted in the best characteristics.

3.3. CONCLUSIONS

In this chapter the fabrication of detectors (diodes) on HP-Si has been discussed. The challenge has been to find that process which produces detectors with the lowest leakage current.

First, the leakage current and related topics were described theoretically. This current is mainly due to the collection of thermally generated carriers. A distinction is made between the depleted and the undepleted part of the diode, as in the former the charge is collected by drift, while in the latter it is collected by diffusion. Moreover, the generation, characterized by the lifetime, is different in both regions. As the temperature dependence is different for both components, they can be distinguished from each other by

51

* * * * * * GRAPHICS PLOT * * * * * * IS 498 2 DI

1.000 /div

.0000

a) VB 1 0 . 0 0 / d l v ( V)

x x x x x x GRAPHICS PLOT x x x x x x IS 498 2 I

(ni)

1.000 /div

.0000

/ /

/

b) va 10.00/dlv t v)

x x x x x x GRAPHICS PLOT x x x x x x IS 496 2 D

xxxxxx GRAPHICS PLOT xxxxxx IS 498 9 I

1.000 /aiv

.0000

/ /

a. ooo /div

.0000 '

^ S S

c)

10 M)

-80.00

x x x x x x GRAPHICS PLOT x x x x x x IS 498 4 I

e) VB 1 0 . 0 0 / d l v ( V)

Fig.3.2.6. IV -characteristic of diodes made by combinations of processes I and III; a), b) and c) hi-junction made by both processes (IS498-2), while the pn-junction is made by a) both processes, b) process III, and c) process I; b), d) and e) pn-junction made by process III, while the hi-junction is made by b) both processes (IS498-2), d) process III (IS498-5), and e) process I (IS498-4).

52

measuring the leakage current's temperature dependence. Secondly, the measurement results obtained on some test diodes made on

HP-Si were presented. In principle, their structure was identical. However, their processing was not. It appeared that the leakage current is greatly dependent on the process used. Implantation together with an annealing at 600 °C, as proposed by Kemmer [3.32], resulted in the best devices, whereas the worst resulted from an implantation and a 900 °C anneal (which is advisable for the electronics processing). The diffused samples yielded an intermediate case. However, diodes made by the combination of diffusion followed by implantation revealed the best characteristics. Measurement of the activation energy showed the presence of a diffusion component in the leakage current which explained the steep increase at the start of the /F-characteristic.

The effect of combined processes has also been investigated. Such combinations can be useful for integrating electronics circuitry with the detector. It was found that the sequence in which the different steps are applied is of major influence on the final detector characteristic. When a high temperature step is applied before the implantation of the detector junctions the characteristics appeared to be comparable, whereas in the other way around they were severely deteriorated. Therefore, the detector junctions have to be implanted after the very last high-temperature step.

CHAPTER 4 OPERATION

4.0. INTRODUCTION

In this chapter two aspects concerning the operation of a strip detector will be discussed. Part I, Sec.4.1., deals with the pulses induced on the detector's strips during charge collection, while Part II, Sec.4.2., considers thé influence of the strip width on the performance of the detector.

. When a particle hits a strip detector, it generates electron-hole (eh) pairs. The collection of these will cause an output pulse at the strips, which enables the particle to be detected; its position is indicated by the position of the collecting strip(s), see Chapter 2.

During their movement towards the strips the generated charge carriers will induce charges in all strips. At the end of the whole collection process the total amount of charge induced in a strip will equal the amount of charge collected by that strip [4.1]. The pulse responses on both kinds of strips (i.e. strips which are or those which are not going to collect the generated charge) have been calculated [4.1], using a new method based upon Tellegen's theorem [4.2].

The pitch and the strip width are the main parameters characterizing a strip detector. Usually, the strip geometry is designed symmetrically, with the strip and gap widths equaling each other. However, a variation of these may influence the performance of the strip detector. Therefore, the effect of such a variation has been studied, both theoretically and experimentally [4.3]. Firstly, the potential and electric field distributions in the neighborhood of the strips were calculated, which enabled the influence of parameters such as detector geometry, bias voltage and surface charge to be investigated. Secondly, test devices were designed and fabricated. These devices, in which both the pitch and the strip width were varied, were tested with respect to their electrical behavior as well as their interaction with radiation.

As a strip detector is intended to be used for position measurement, before taking a further look of these two aspects, a short discussion will be given in the next subsection on the position precision of a strip detector.

4.0.1. POSITION PRECISION

As minimum ionizing particles travel with the speed of light, they will pass a detector in a few picoseconds, leaving behind a 1 jjm wide tube of generated electron-hole (eh) pairs [4.4,4.5] (Sec.2.2.1.). In a fully-depleted detector these will be collected in typically 10 ns [4.4,4.6]. The carriers are not generated in a high density, so that space-charge (plasma) effects do not have to be accounted for [4.7,4.8,4.9].

Due to so-called high-energetic 5-electrons the center of mass of the

54

generated charge tube can shift; the chance of a 1 /jm shift is about 18% in the case of 300 /mi-thick detectors [4.7]. In the case of angled tracks the shift will be larger, as the energy deposition is not homogeneously divided along the particle's track [4.7,4.8] (For a 300 /mi long path the standard deviation of the center of gravity along that path will be about 12/im [4.8]). For thinner detectors this value will, of course, be smaller. By rejecting events in which a high energy deposit has occurred (i.e. events with a relatively large amount of 5-electrons), a smaller spread will be achieved [4.8]. When a magnetic field is present a systematic shift in the position coordinate will occur [4.5,4.8]. Due to the mounting precision an additional uncertainty of about 1 fim will result [4.4].

During the collection the tube of charge carriers widens due to diffusion. (Depending on its density electrostatic repulsion has to be accounted for as well [4.10], which in the case of minimum ionizing particles is not required for the microstrip detector [4.5].) Typically, 8 pm broadening results during a 10 ns collection [4.4,4.6,4.7,4.11]. Consequently, depending on the strip pitch the charge will be collected by more than one strip (thereby setting a minimum utilization on the strip pitch; see also Sec.4.2.1.). As the carriers generated at the rear of the detector have to be collected over a larger distance than those generated at the front, they will also diffuse out over a larger area. A complex description of the final distribution of the collected carriers will result [4.5].

The effect will be that, when the position of the incident radiation is varied in the direction perpendicular to the strips, the total amount of collected charge will decrease very rapidly at the collection strip, while simultaneously increasing very rapidly at the next one [4.5]. It will not linearly decrease on one and increase on the other. So, there are more or less two ways to measure the position with a strip detector (provided that the strip pitch is on the same order as the width of the outdiffused charge tube): digital or analog. In the first case the position is determined by the strip with the largest output pulse and a staircase-like input-ouput function results. The noise level of the detector is less crucial and the position precision is equal to the strip pitch. In the second case the position is determined by weighting the output pulses from some adjacent strips, resulting in a more gradual input-output function. A low noise level is preferable and more signal processing is required, but a better position precision can be obtained [4.6]. (In this way a position precision of about 3 urn is achieved with 20 nm pitch detectors [4.4,4.5]. It should even be possible to determine the center of mass with a 1 urn precision with these detectors [4.8].)

In principle, a strip detector is a digital position detector, i.e. the position indication function is a square (top-hat) distribution (especially in the case of a large pitch with wide strips). Due to the outdiffusion during collection and the noise from the electronics, etc. a Gaussian distribution will be added to this one. Usually [4.12], a standard deviation a is assigned to the initial top-hat distribution f(x) by

55

° „OO ff(x)&x pJ~ip J-oo

Uxtdx"-12' ( 4 - ° - J )

where p denotes the strip pitch. This value will be measured when microstrip detectors with a relatively large pitch are used. However, in the case of a smaller pitch, making weighting possible, a much lower value will be found [4.4,4.8].

Placing a few detectors one after another should increase the position precision, as the particle will traverse them all. In the case of detectors with a Gaussian distribution this is simply true; an improvement of a factor of VN will be reached with N detectors [4.12]. However, in the case of digital detectors the result depends on the way the detectors are stacked [4.12]: exact alignment (strip above strip) will not give an improvement, whereas a uniform shift will give an improvement of N (using N detectors).

4.1. PARTI - THE PULSE RESPONSES ON A DETECTOR'S STRIPS

4.1.0. Introduction

When a particle traverses a strip detector, it generates electron-hole pairs. The collection of these causes an output pulse at the detector's electrodes, enabling the particle to be detected. During their motion towards the electrodes, the generated charge carriers induce a current in the electrodes in such a way that, upon collection, the total amount of charge emitted by each electrode is equal to the total amount of charge collected by it. After that moment, equilibrium is established again.

The charge induced in an electrode can be estimated by considering the instantaneous change of electrostatic flux lines which end on that electrode due to the movement of the charge carrier [4.13]. For a detector with only two electrodes the shape of the induced pulse can be calculated by Ramo's theorem [4.13,4.14]. However, in the presence of more electrodes, as in the case of a microstrip detector, only the overall response of all the electrodes together can be calculated in this way [4.14]. In order to find the pulse on a certain electrode, a different method has to be applied. This case is of particular interest, because of what happens to the induction on two neighboring strips. This induction should be the same on both when the moving charge is still in the deep bulk. However, when the charge is going to be collected by one of the strips the total amount of induced charge on that strip will eventually equal this charge, whereas on the other strip this amount has to become zero. Consequently, the induced pulse on that strip will become negative.

Formulas that can be used to calculate these induced pulses have been presented [4.15,4.16]. However, a more general expression has been derived

56

[4.1]. Based upon Tellegen's theorem, known from network analysis [4.2], it is shown that the current induced in a detector strip can be calculated by scaling the currents flowing inside the detector (resulting from the movement of the generated charges) by a reference field and then integrating these weighted currents. This field - determined separately - depends solely on the geometry of the detector. Its generality makes the method applicable to other induction problems as well.

First, in the next section the theory by which the induced pulses can be calculated is outlined. In the section thereafter the results of such calculations will be presented.

4.1.1. Theory

The shape of the induced pulses is analyzed with the aid of Tellegen's theorem [4.2]. This theorem is described in the next subsection, and its application to compute the desired pulses is discussed in the following one.

Table 4.1.1. List of symbols.

<f> -E -D -J -V -I -V -s -n_ -P -Q ~

! -r_ t -e A» -

electric potential electric field strength electric flux density (= £_E) volume density of electric current electric potential electric current domain surface unit normal (directed away from V) volume density of electric charge elementary charge velocity (= /z_E) position vector time permittivity mobility

4.1.1.1. The equations

Let us consider a bounded domain V with closed boundary surface S (Fig.4.1.1.). The electroded parts of S are denoted by S0,S1,...,SN. These have the constant electric potentials V0,V1,...,VN, respectively, and carry the electric currents V0,/!,...,7N away from V (Fig.4.1.1.). Next, define two possible distributions of quasi-stationary currents and fields (labeled A and B) in V.

57

Fig.4.1.1. Bounded domain V with closed boundary surface S, where S0,SV...,SN are the electrodes.

The field with label A satisfies the equation (see Table 4.1.1. for the meaning of the symbols)

V x ^ = 0 in V. (4.1.1)

Hence, an electric potential <j> can be defined such that

ff* = -V<f>A mV.

At the electrodes we have

<t>A = V$ a t S n (n = 0,l,...,N).

The field with label B satisfies the continuity equation

V.(J? + dtDB) = 0 inK.

(4.1.2)

(4.1.3)

(4.1.4)

Further, the current carried away from V through the electrode Sn is given by

/ ? - (ZB + dtDB).n_dA (n = 0,l,...,N). (4.1.5a)

At the remaining, i.e. insulating, part of 5 we have

58

r.n = o. (4.1.5b)

Then, consider the expression

^ • ( Z B + dtMB) = - / . ( j f + dtDB)

= - v . [ A z B + 3 toB)] + <t>K v.(ZB + atDB) A . ,B , „ B , = -V.[A/ + 3/)], (4.1.6)

where Eqs.(4.1.2) and (4.1.4) have been used. Next, integrate Eq.(4.1.6) over the domain F and use Gauss's theorem [4.17]; the result is:

£*.(/* + dt_DB)dV = - ( ) n_.^A(_JB + a t D B ) d A V J S

= -t ^/B Z-i ' n ' n

n=0

S-(5'0+51+...+Si,) n ^ A d t D B d A , (4.1.7)

where Eqs.(4.1.3), (4.1.5a) and (4.1.5b) have been used. Next, use the relation D = eZT to rewrite parts of Eq.(4.1.7) as follows:

£ A . d t .D B dV = _DA.at£;B dV = D A . - V d t 0 B d V

■V.(Z)A 3 t ^ B ) d V + (V.Z)A)d t <6BdV

= ' ^ ü - ( ^ A 3 t ^ B ) d A + 5

/ a t ^ B d V , (4.1.8)

A A where Eq.(4.1.2) and the relation _V._D -p have been used.

Finally, by combining Eqs.(4.1.7) and (4.1.8), Tellegen's theorem [4.2] (in a generalized form) results:

59

Etf '„ B - E^.J* dV n = 0

N

n=OJ n_.(DA dt(f>B) dA - P A a t^B dV

S-(S0+S1+...+SN) ni.(DA a t ^ B - / 8tDB)dA. (4.1.9)

Eq.(4.1.9) enables the pulse response on a strip of a strip detector to be calculated: In state A all strips are set to ground potential, except for the one, say strip m, whose pulse response has to be calculated; that one is set to 1 V, i.e.

0 if n # m,

1 if n = m. (4.1.10)

Further, we take p =0. In state B the actual detector is represented (bias conditions, space charge

and so on), in which eh pairs are generated, resulting in a current

ZB(z) = E kl/^Wto) - E«(i)*(£). (4.1.11)

where q is the charge of the carrier, _v its velocity and S(_r_) the three-dimensional Dirac function. Further, it is supposed that all strips are kept at a constant potential and hence that

dt4> = 0 a t 5 n (n = 0,l,...,N). (4.1.12)

By a more refined analysis in which the vacuum around V, extending to infinity, is observed as well, it can be shown that the last term in Eq.(4.1.9) vanishes. Then from Eq.(4.1.9), the value of the current flowing out of strip m during the collection of some generated charge (represented by J_ ) is obtained as

/ B = - E^.fdV. (4.1.13)

60

4.1.1.2. Discussion

In fact, Eq.(4.1.9) represents a generalized form of earlier versions [4.14,4.15]. Ref.[4.14] arrived at

n=0 E^.^dV, (4.1.14)

in which state A is identical to state B except that no space charge is present in state A. The above analysis showed that the effect of the presence of this space charge is an extra term to be evaluated (Eq.(4 1.9)), which, of course, compensates for the change in the electric field j± brought about by this space charge.

Ref.[4.15] has derived for the current 7m through electrode m, and due to a charge q moving with velocity v_ the expression

£ = - « v . - p r . (4.1.15)

The last factor represents the change in the electric field caused by a small variation in the potential of electrode m; all the other electrodes are kept at a constant potential. Eq.(4.1.15) can also be found from Eq.(4.1.9) by taking two identical cases in which only Vm differs slightly and by subsequently subtracting the two from each other, resulting in

/ B = m V

<9 /" -=r.JBdV. (4.1.16) dVZ ~

By substituting Eq.(4.1.11) in Eq.(4.1.16), Eq.(4.1.15) results. In fact, the result is even more general, since states A and B are chosen independently (only the geometries have to be the same). As a consequence the factor 9E^/dVm should be the same for every configuration (e.g. vacuum as well as space charge). The geometry is the only determining factor. Upon comparing Eqs.(4.1.13) and (4.1.16), it can also be observed that when no space charge is present that

8E E

-w- =T- ( 4 - L 1 7 > (Keep in mind that in the derivation of Eq.(4.1.13), V^ was set to 1 V). This result has been mentioned and discussed earlier [4.15].

Ref.[4.16] approached the problem differently. They studied the case in

61

which the electrodes are floating or grounded via resistors. So, contrary to the former approaches [4.13,4.14,4.15] as well as this approach [4.1], Eq.(4.1.12) doesn't hold. By applying Green's reciprocity theorem [4.17] they showed that the voltage induced in an electrode (or current in the case of a grounded electrode) by a moving point charge can be calculated by the time convolution of a weighting field and the current density representing the moving charge. The weighting field is found as the time response of the electric field due to a unit delta pulse of current (or voltage in the case of a grounded electrode) applied at the output electrode of interest, while no space charge is present in the system. In the case in which all electrodes are grounded this response will also be a delta pulse and the convolution simplifies to an expression similar to Eq.(4.1.13) [4.16]. In this approach the weighting field is determined independently, as it is in the one presented (see below).

Eq.(4.1.13) can be used to calculate the current from a detector strip when generated eh pairs are being collected. In view of Tellegen's theorem, this pulse response is built up from two independent parameters: the field Er and the current J_ . Here, Ey is solely determined by the geometry of the detector, whereas J_ is also determined by the applied bias, the space charge density and the junction site. Eq.(4.1.11) implies that the total output current can be found by summing the individual currents arising from each separate generated electron and hole:

£ ( 0 = -E^-«v(r(/)). (4.1.18)

Since E* and J_ are independently defined, it can be shown a priori that the final amount of charge induced in a strip will equal the total amount of charge collected by that strip (as it should). Suppose that S0 (Fig.4.1.1.) represents the back side of the detector and 5'1,...,5'N are the strips. Then, by integrating the field Ey from the back side to a strip, one obtains (note the definition of state A, Eq.(4.1.10)):

f

~"_A _A .A 0 if n # m, E^.dr. = Vt-Vy = (4.1.19)

-1 if n = m. -10

Suppose that all the charge is generated at Z=J!g> a n d that the holes (of total charge +Q) are collected by strip k (and thus the electrons, of total charge -Q, by the rear contact). From this it is clear that

'coll _ . - . R 0 if m # k,

tdt = (4.1.20) -) Q if m = k,

62

should hold. Here, /coll is the total time needed to collect all of the generated charge, i.e.:

'coll.e

£o

and/ c o l l = max(/co l l , / c o l l h ) .

'coll.h **

-Lk *Hr)dr,

£ g

(4.1.21)

The current flowing in the detector at time / is

h e

Hence, with Eq.(4.1.13) the output pulse on strip m is

e h

Next, by using Eq.(4.1.19) it follows that fcoll

G > = E<7

= E<?

coll.e

^. j fd* - E<? f'coll.h

J/1 df

.10

tf\d£ - Y.9 h

dr

= -E? E ^ d r = 0 if m # k ,

Ö if m = k ,

(4.1.22)

(4.1.23)

(4.1.24)

which is in accordance with the expectation expressed by Eq.(4.1.20).

4.1.2. Results

With the theory presented in Sec.4.1.1., culminating in Eq.(4.1.13), the output current pulse from a strip-detector strip was calculated when, somewhere in the detector, generated charge was being collected. The integration in Eq.(4.1.13) is replaced by a summation (cf. Eq.(4.1.18)), due to the point-like character of the charge carriers. Below the model on which such a calculation has been based is outlined first, and some results are presented after that.

63

4.1.2.1. The model

According to the theory of Sec.4.1.1. the field and the current J can be calculated separately. The current J_ is known from the actual electric field E_ in the detector (Eq.(4.1.11)), assuming that no saturation effects occur (Sec.4.2.1.). In the major part of the detector the electric field _£ can be determined by using one-dimensional theory (Sec.4.2., [4.3]). In the calculations, therefore, the current J_ was determined by assuming the validity of one-dimensional theory throughout the whole detector. It is expected that this assumption will not significantly change the computed results. For the same reason, the ever-present lateral outdiffusion of the generated charge carriers [4.5,4.11] is also neglected in the calculations.

By virtue of both the dot product and the assumed direction of J_ , only the component of E_ in the y-direction needs to be calculated (Fig.4.1.2., Eq.(4.1.13)). The strips are assumed to be of infinite length, thereby reducing the problem to a two-dimensional one. In order to calculate the field Ey, first the potential distribution is calculated. As stated before, in state A no space charge is present. All strips, the rear contact included, are at zero potential, except the one whose induction pulse has to be known. That one is set to 1 V, see Fig.4.1.2.

nu„

Fig.4.1.2. Frame of the model.

In the calculations the strip pitch was chosen to be 20 fxm. The detector's thickness was chosen as 350 and 450 pm. As it appeared that this parameter did not have much influence on the results, only those obtained with the 350/zm thickness are presented. The detector width was chosen so large that no errors were introduced by setting the boundary between the front and rear sides of the detector to 0 V (state A). For reasons of symmetry only one half of the detector is needed for the model (Fig.4.1.2.). Along the boundary formed by the symmetry axis, the potential was set to vary linearly between

64

1 V and 0 V (as it was as well between the first strip and its neighbor, Fig.4.1.2.).

With these boundary conditions the potential distribution (state A) was calculated by solving the two-dimensional Poisson equation with the aid of a finite difference method with successive over taxa t ion [4.18]; the result is shown in Fig.4.1.3a. Finally, the y-component of the electric field Ey is calculated from this distribution, with the result as shown in Fig.4.1.3b.

a)

Fig.4.1.3. a) Potential distribution <f> and b) the resulting y-component of the field E . 500 pm from the first strip onwards is shown.

Now, since J and are known, the pulse responses at the different strips can be calculated (Eqs.(4.1.13) and (4.1.18)). This is done simultaneously by varying the x-position of the current J_ while using the same distribution of E .

65

In the calculations six different generation sources were considered: five point-like sources and one line-like source. The latter represents, for example, a minimum ionizing particle. The former represent a part of the total generated charge of a line-like source (e.g. minimum ionizing particles entering the detector slantwise), and show how the pulse from such a source is composed. They were positioned either at the surfaces of the detector (then also representing, for example, heavy ions or low energy particles), in the middle or at a depth of a quarter or three quarters. The total generated charge was in all six cases normalized to 1 fC.

For the field E_ inside the detector, through which the current density (and the collection times) is determined (Eq.(4.1.11)), there were again six possibilities: a constant field (representing no (net) space charge, strong overdepletion, or saturation) in both directions, and a linearly varying field of the four possibilities of n- or p-type bulk with the junction at the front or the rear side. Because electrons and holes have different drift velocities, both polarizations of the electric field were investigated. The constant field was chosen to be 30V/100^m. In the other four cases a doping density of 10 cm" was selected, the detector being 1% overdepleted.

For completeness, the value of the elementary charge was set to 1.6 10 C, the permittivity of silicon e was chosen as 1.04 10 F/cm the electron mobility fin as 1350 cm /Vs and the hole mobilty fip as 480 cm /Vs [4.19].

4.1.2.2. The pulses

The pulses resulting from the computation were in accordance with previously published data [4.13,4.14,4.15,4.20]. The most significant ones are shown in Figs.4.1.4.-4.1.10. The position of the generation source (the impinged particle) perpendicular to the strips (the x-direction, Fig.4.1.2.) is taken as a parameter in these figures. The sign of the pulses is such that the current flow out of the readout strip is shown. So, for p-type strips (negative

Table 4.1.2. Legend to Figs.4.1.4.-4.1.10.

symbol

X o + o A □

generation source x- position

(/urn)

0 10

20 50 90

125

notes

collection by readout strip collection by readout strip

and its next neighbor collection by neighboring strip(s)

ti

it

ii

66

field; collection of holes) a positive sign will result. The pulses are plotted for six x-positions of the generation source, see Table 4.1.2. The vertical scale goes from -200 to +200 nA in every plot; the horizontal scale, however, differs per plot and depends on the collection time, which was usually about 25 to 72 ns. The pulse was integrated for every position. As predicted by Eqs.(4.1.20) and (4.1.24) the result was 1 fC (collection of holes) or -1 fC (collection of electrons) at 0 /mi, 0.5 fC or -0.5 fC at 10 nm and 0 C elsewhere.

Because of Eq.(4.1.12), the results are based upon the assumption that the strips remain at a constant potential during the charge-collection process. When a strip is floating, however, its potential will of course vary in a way something like the following (exact expressions are derived by [4.16], see Sec.4.1.1.2.):

AK = -^AÖmd = ^ ' indA' , (4-1-25)

in which Cs represents the strip capacitance. When the amplifier connected to a strip has an input resistance R, the induced current will be divided into two parts: one part flowing into the amplifier, thereby introducing a potential rise due to the input resistance, and one part charging the strip capacitance according to Eq.(4.1.25), causing the same potential rise in the strip potential. At this stage it should be noted that, when the integration time constant of the amplifier is larger than the collection time, the effect of the induction will have been filtered at the output of the amplifier, and only the real amount of charge collected by the strip will be seen at that output node (unless the input impedance of the preamplifiers is larger than the interstrip impedance, which causes crosstalk).

Once again it is stated that diffusion effects are neglected. This means that the widening of the tube of the carriers generated during the collection process [4.5,4.11] is not accounted for; nor is the collection by diffusion. Only collection by drift is taken into account, and that according to one-dimensional theory. Consequently, results obtained with the generation source located at the surface between the readout strip and its neighbor have to be considered suspect (Sec.4.2., [4.3]), as only the induction by one carrier (electron or hole) is computed, the other having been considered as having been collected already.

Fig.4.1.4. plots the pulse responses for a point-like generation source and a constant negative electric field (-30V/100/xm, collection of holes; in the case of a positive electric field the results are analogous). The generation source is at respectively the surface, a quarter depth, in the middle and at the rear surface of the detector. By virtue of the constant electric field, and therefore the constant collection current, the pulses will reflect the shape of the reference field As the generated carriers traverse the detector in the opposite direction, Fig.4.1.4d. is the mirror image of Fig.4.1.4a.

67

6 8 10 12 time Ins! ■-

9 12 15 18 21 24 27 30 time (ns) »-

d) c) Fig.4.1.4. Pulse responses for a constant electric field (-30V/lOOpm, collection of holes); the point-like generation source is at a) the surface, b) a quarter depth, c) in the middle and d) the rear surface of the detector.

As the mobility of electrons is three times higher than that of holes, the electrons will induce a current three times as high, albeit of shorter duration (cf. Figs.4.1.4a. and 4.1.4d.). This is clearly demonstrated in Fig.4.1.4c, where at one third of the total collection time the induced current drops to one quarter of its value, indicating that the electrons are collected. In Fig.4.1.4b. the electrons and holes are collected in the same time (the collection time is therefore the shortest in this case), resulting in the induced pulse having a negative part at both the start and the end of the pulse.

Figs.4.1.5.-4.1.8. illustrate pulse responses for a point-like generation source and a linearly increasing or decreasing field (depending on the site of the junction) for both n and p-type bulk material. Due to the linearly varying field the velocity of the charge carriers is dependent on their position, thereby affecting the course of time. For example, when the junction is at the front, the velocity of the carriers will be the largest in that area where the field is also the strongest (Fig.4.1.3b.), causing a short, strong negative compensation dip to arise (Figs.4.1.5. and 4.1.6.); or the

68

O 8 16 ït, 32 M) 4B 56 64 72 time (ns) *-

c) Fig.4.1.5. Pulse responses for an n-type bulk with the junction on the front (strip) side. The point-like generation source is at a) the front surface, b) the middle and c) the rear surface.

Fig.4.1.6. Pulse responses for a p-type bulk with the junction on the front (strip) side. The point-like generation source is at a) the front surface and b) the rear surface.

69

O 6 16 24 32 40 48 56 64 72 time (nsl •-

a)

Fig.4.1.7. Pulse responses for an n-type bulk with the junction on the rear. The point-like generation source is at a) the front surface, b) the middle and c) the rear surface.

16 21 32 40 1.8 56 64 72 time (nsl -

*; Fig.4.1.8. Pulse responses for a p-type bulk with the junction on the rear. The point-like generation source is at a) the front surface and b) the rear surface.

70

-a o

Fig.4.1.9. Pulse responses for a line-like generation source. The collection field is constant 30V/100p.m: a) positive field, collection of electrons and b) negative field, collection of holes.

0 8 16 24 32 1.0 48 56 64 72 time Ins) »-

b)

d) IK 32 U0 i.8 56 6*.

time (ns) *-

c) Fig.4.1.10. Pulse responses for a line-like generation source, a) n-type bulk, junction at the front, b) p-type bulk, junction at the front, c) n-type bulk, junction at the rear and d) p-type bulk, junction at the rear.

71

contrary, in the other case, the negative part of the induced pulse will not be that pronounced (Figs.4.1.7. and 4.1.8.).

In the case of Fig.4.1.5b. the holes, giving rise to a reasonable signal, are collected first. The remaining electrons have such a low velocity that, coupled with the field size Ey, only a small induction remains until they are also collected.

When the junction is at the rear (the strips are still at the front), the largest currents J_ do not coincide with the largest field £r \ So, the resulting induced pulses can only be predicted poorly, and capricious curves arise, Figs.4.1.7. and 4.1.8., especially when the charge carriers have been generated somewhere in the bulk, causing the flow of both electrons and holes. (As electrons have the higher mobility, the most curious curves will then be seen in the case of an n-type bulk.)

Figs.4.1.9. and 4.1.10. depict the pulse responses when the generation source is of a line form, the charge being homogeneously generated, as in the case of, for example, minimum ionizing particles. In the calculations, every mesh point along the line (situated at every 10 fim) was assigned an equal amount of charge at / = 0. Due to this discretization of the source, spike or staircase-like irregularities arose in the pulse responses in the case of a varying field E_ (Fig.4.1.10.).

In the case of a constant electric field, the electrons will be collected at one third of the total collection time, as can clearly be seen in the curvature of the responses, Fig.4.1.9. (This is even the point where the induced pulses change their sign.) Likewise, in the case of a linearly varying field, all the charge will be collected in 72 ns (given the particular settings), but, as can be seen from Fig.4.1.10., the signal has nearly vanished after about 24 ns, when all the electrons are collected and an "enormous" tail of 48 ns is left. This is the most clearly evident, when the junction is on the front side (the largest field ET coincides with the largest current J_ ), or when the electrons (having the higher mobility and thus a dominant contribution to J_ ) are collected by the strips.

Compared with the case of a point-like generation source (e.g. heavy ions or low energy particles), for a line-like generation source (e.g. minimum ionizing particles) the induced pulse on a strip next to the one which is going to collect the generated charge is much smaller. This effect is the strongest (i.e. the induction is the weakest) when the junction is on the strip side. The effect is the weakest for a p-type bulk with p-type strips, see Fig.4.1.10.

4.2. PARTÜ - THE INFLUENCE OF THE STRIP WIDTH

4.2.0. Introduction

In designing a strip detector, it is usual to choose the strip width arbitrarily to equal half the pitch size. In order to gain more insight into the functioning of a strip detector, especially the influence of this strip width on the detector performance, a study has been made of the influence of this strip

72

/-Si02 /-surface charge /-field plate

~" '///P-—W//////r^ V//\

n- Si

a)

geometry [4.3]. This was carried out theoretically by numerical calculation of the potential and electric field distributions in the neighborhood of the strips, and experimentally by measurements on fabricated test structures. The results will be discussed in the following sections.

In Sec.4.2.1. the numerical model will be stated and discussed. Some minor results obtained by the calculations are also included in that section. The influence of more important parameters are discussed in the subsequent sections, namely the influence of surface charge (Sec.4.2.2.), of field plates (Sec.4.2.3.), and of the bias voltage (Sec.4.2.4.). In Sec.4.2.5. the influence of disturbances on the strip potential with respect to the strip geometry is discussed. Lastly, in Sec.4.2.6. the results of experiments on the fabricated detectors are presented and discussed.

r »r

73

LEGEND:

1

2

3

4

5

6

7

3 103

1 103

3 102

0 V/cm

- 3 102

-1 103

-3 103

d)

c) Fig.4.2.1. a) The model for the strip detector, b) A typical potential distribution, c) A typical distribution of the electric field Ex. d) Legend for the electric field distributions. The dimension of the mesh is 1 p.m x 1 \im. See the text for the other data.

4.2.1. Numerical model

By calculating the electric fields in a strip detector a better insight into its functioning can be achieved, especially with respect to the influence of the strip geometry, of disturbance voltages on those strips and of effects like surface charge. Therefore, a computer program was developed to calculate these fields [4.3], As the strip length is usually much larger than the strip pitch, a two-dimensional model will suffice for this calculation (Fig.4.2.1 a.).

Strip detectors are usually made in n-type high-purity silicon (as were the test structures, Sec.4.2.6.). Therefore, in the model the bulk was also assumed to be of n-type silicon. Of course, in the case of p-type bulk the results will be analogous. In general, it is only important to know the type for sign definitions. An exception concerns the effect of surface charge (see Sec.4.2.2.), where there is some difference, because the oxide charge is normally positive.

The bulk doping concentration, assumed to be homogeneously distributed, was mostly set to 10 atoms/cm . The geometry and the amount of strips

74

could be freely chosen. The junction depth was set to 3 jzm, as that is the depth which will result in this high-purity silicon when the detector is made by the diffusion process (Table 3.2.1.). For the thickness of the oxide layer 0.4 /zm was chosen. Albeit arbitrary, it is a typical value of this layer's thickness. Finally, surface charge at the silicon - silicon dioxide interface (between the strips), and field plates on the oxide layer, having some fixed potential, could be incorporated in the model (Fig.4.2.1a.).

The potential of these field plates was directly used as the boundary condition at these places, as were the potentials of the strips (mostly 0 V) on the strip spots (the shaded areas in Fig.4.2.1a.). In order to achieve boundary conditions at the remaining places on the surface, a fixed potential of .0 V was defined at an imaginary plane in the vacuum far above (1 mm) the surface. At the bulk contact the bias voltage (usually 100 V) was used to define the boundary conditions, while in the x-direction the periodicity of the structure was used for this (Fig.4.2.1a.). The permittivity changes at the Si02-interfaces are taken into account as they ought to be [4.17].

The field pattern was calculated by first solving (only) the two-dimensional Poisson equation with the above-mentioned boundary conditions, yielding the potential distribution in the defined area. (The depletion approximation [4.21] has been assumed.) A finite-difference numerical method was used (usually the underlying mesh was 1 x 1 /zm2). Through this method the Poisson equation was transformed into a set of equations which could then be solved iteratively, using SOR (successive over relaxation) [4.18]. The one-dimensional potential distribution (cf. Eq.(4.2.1)) was used as the starting approximation. The depletion approximation was incorporated by assuming all the dopant atoms to be ionized. When during the iteration process a potential lower than 0 V or higher than the applied bias voltage (the potentials at the contacts) was calculated, it was directly set to respectively 0 V or the bias voltage. The appearance of such a potential implies the existence of a local minimum for holes or electrons, respectively. Free carriers, which by virtue of thermal generation are always present in the device, will migrate to these minima, and thereby readjust the potential until the potential at the contacts is finally achieved [4.22]. In this way it is not necessary to know the border of the depletion layer in advance; it will be determined by the program.

The potential growth in (the depleted part of) the strips themselves is neglected, as this quantity will be very low, due to the large difference in the doping concentrations in the strip and bulk areas (cf. [4.21,4.23]).

Once the potential distribution was found, the electric fields could be calculated, based upon this distribution and the known difference mesh used to solve the Poisson equation. In fact, only the separating field Ex in the x-direction is of interest, as this field is responsible for the separation of the charge carriers to the strips, and therefore for the position resolution of the detector. The collecting field Ey in the y-direction is responsible for the transport of the charge carriers to the surface and is already known from one-dimensional theory [4.21,4.23].

75

In Figs.4.2.1b. and 4.2.1c. examples of the potential distribution and the electric field in the x-direction are depicted as they will be shown throughout this section (Sec.4.2.). Only the silicon part is shown, as the part in the oxide layer (0.4 p.m in thickness) is suppressed. In this example, there is no surface charge or field plate. The bias voltage is 100 V. The strips, having a pitch of 20 /mi, are 10 fim wide. In the plot of the potential distribution, the potential is plotted in a three-dimensional view, whereas the field is depicted as contour lines; the legend numbers are listed in Fig.4.2.1d. As holes will be collected by the strips, the first plot clearly visualizes how these holes will move downwards to those strips, while the second provides more exact information about the separation process.

During the calculations it appeared that when the thickness of the oxide layer was varied, the field distribution did not change noticeably. Varying the junction depth did not have much influence on the actual field pattern either, i.e. the pattern from this depth onwards into the substrate. The separating power remained almost unaffected. The effect of deeper junctions will be a diminished influence of the Si-Si02 interface, but, on the other hand, an increasing difference in dead layers at the strip and interstrip regions. Moreover, the interstrip capacitance will be increased. With respect to the results from these numerical calculations, therefore, no serious difference is to be expected between junction-type detectors and their surface barrier equivalents, as in the model the latter will be like the former with no junction depth.

The calculations made it clear that the potential distributions are in fact determined by the (interstrip) gap. The pitch is of no importance. The gap is the cause of the separating field E^, and is therefore responsible for the position sensitivity of the detector. Modeling different pitches, but maintaining the same gap widths resulted in the same distributions (in the

p z a ? Y//////A ¥ V/////A 1 Y//////X 1 r7773 ( l i l i ; i i i i / I ; i i i \

( 1 - dimensional solution / I no position resolution /

'////////////////////////////////////////////////////////, Fig.4.2.2. The effect of the strip pattern is manifested one gap length into the substrate.

76

neighborhood of the gaps). A strip detector can therefore better be seen as one large plane diode with disturbances (the gaps) in it, rather than as a collection of strip-like diodes, put together in a row.

About one gap length into the substrate the one-dimensional solution was retrieved (Fig.4.2.2.). Therefore, the distributions will be shown to only this depth (cf. Fig.4.2.1.). Although in principle the bulk thus has no position resolution, the effect of the strip pattern will be as if the detector's collection volume is split up into compartments, each belonging to one strip. These compartments have the width of one pitch length. Consequently, the detector's position precision is dictated by this pitch. On the other hand, because there is no field in the x-direction in the bulk, the charge carriers can diffuse out in that direction during their drifting towards the strips. At the surface they can then be directed to different strips, and a more precise position estimate can be made by pulse weighting [4.5] (see also Sec.4.0.1.). (It is assumed that the carriers move according to the electric-field pattern. The £^=0-Iine therefore indicates the border between the different compartments. Still, due to their thermal velocity the carriers can be collected by the other strip, thereby giving rise to a kind of "partition noise". However, being balanced, the effect is assumed to be negligible.)

The choice of the gap width does not influence the position precision, but it does influence the separating field Ex, and therefore the electrical properties of the detector, as will be seen in the following sections. Secondary effects, which can appear at large widths, are pulse delay or broadening due to an increased collection path by which the charge carriers have to be collected (the field E^ is not strong enough so that this effect can be neglected a priori), and a more prominent influence of the Si-Si02 interface (surface leakage current, trapping and so on). The separating field £x is smaller at smaller gap widths, but then the required deflection is also smaller. However, at extremely small widths, the detector's functioning as a position detector will become questionable. The field Ex is too small, so that the detector will behave as a single plane diode with only strip-like contacts, which has no position resolution.

The maximum electric field will be at the edges of the strips. The calculations revealed values around 10 V/cm for a bias of 100 V (slightly more than one-dimensional theory would indicate [4.21,4.23]). This indicates that effects such as multiplication and breakdown are not likely to happen in strip detectors, as they require a field on the order of 10 V/cm [4.21,4.23]. Moreover, the velocity of the generated electrons and holes which are being collected by the detector will remain proportional to the electric field without any saturation occurring.

4.2.2. Surface charge

Due to the Si-Si02 transition there will be surface states at the surface which, depending on the actual surface potential and Fermi-level, can be charged [4.21]. In addition, the ever-present (positively charged) ions in the

77

Fig.4.2.3. a) The potential and b) electric field distribution when surface charge (1x10 cm ) is present. The other conditions are the same as for Figs.4.2.1b. and 4.2.1c.

78

oxide layer are responsible for the presence of surface charge. Finally, by means of implantation the possibility of artificially creating some surface charge exists. It is therefore interesting to investigate the influence of the surface charge on the potential distribution in a strip detector and consequently on its performance as a detector. In the model the surface charge is positioned at the Si-Si02 interface, see Fig.4.2.1a., which allows all of the above-mentioned types of surface charge to be simulated.

Due to the (positive) surface charge the collecting field E will change direction at the surface (Fig.4.2.3.), for this charge controls the field at the surface (cf. Eq.(4.2.9)). This effect only becomes noticeable when a critical threshold is passed. For a bulk doping concentration of 1 x 1 0 cm . this threshold appears to be at 1 -3x10 cm . This value was found for a detector with a 20 /im pitch, a 10 nm gap width, a 0.4 /im oxide thickness and a 3/jm junction depth. However, by varying these parameters no change could be observed in this threshold value.

Due to the surface charge the separating field Ex gets stronger (Fig.4.2.3.; cf. Fig.4.2.1.), resulting in a more pronounced partition of the collection compartments (Fig.4.2.2.). However, a local potential minimum for electrons at the surface between the strips has resulted (Fig.4.2.3.). Following the same reasoning as in the previous section, this bucket will be filled with thermally generated electrons until the minimum has disappeared. (An effective surface charge density of about 1 x 1 0 cm remains.) This means that the effect of the surface charge will be the presence of free carriers, while the potential and electric field distributions will remain nearly unaffected and be like the case without surface charge. (Consequently, most of the distributions presented here are those without surface charge.) The presence of the free carriers will result in an enlargement of the recombination rate at the surface and therefore in a signal loss (only for those carriers generated near the surface; bulk generated carriers will be swept towards a strip before reaching the surface).

The effect is even more clear when the case of negative surface charge (due to, say, a boron implant between the strips) is examined. A potential lower than that of the strips then results, which means that the holes will migrate to those places, thereby compensating the lower potential. The final result will be the existence of a "channel" between the strips, as could directly be deduced from the boron implant. (These channels can be useful when specific resistances between the strips are needed, such as in resistive charge division, Chapter 5.) Because oxide charges are mostly positive, these channels can arise unwanted in the case of p-type bulk detectors with n-type strips [4.24].

In fact, the real structure is three-dimensional, and therefore some care has to be taken with respect to the term "local minimum". When a guard ring surrounds the strips, there will clearly still be a potential minimum, but without a guard ring, there can be an electron flow along the strips out of this potential bucket.

In the case in which the amount of free carriers at the surface must be

79

determined exactly, not only the potential, but also the Fermi level has to be calculated. This means that the current and continuity equations have to be solved simultaneously with the Poisson equation [4.25]. Because the computer program used only solved the Poisson equation, such exact determination could not be carried out.

4.2.3. Field plates

In addition to implanting doping atoms, surface charge can be artificially created by means of field plates. Field plates offer the possibility of changing the surface charge density during operation of the detector by simply changing the potential of the field plate. Also, field plate-like patterns can be present in the metalization masks (e.g. for interconnection of electronic circuitry, or for capacitive multiplexing purposes, Chapter 5). It is therefore interesting to investigate the influence of field plates on the potential distribution and thus on the performance of a strip detector.

The effect of a field plate should be in accordance with the MOS charge induction theory [4.21,4.23]. Depending on its potential it will induce charge at the silicon surface beneath it. Because the computer program only calculated the potential distribution, this charge induction was, like the effect of surface charge, seen as the creation of a local minimum in that distribution (Fig.4.2.4.): When the potential of the field plate is set to that potential which would appear at the silicon surface without a field plate, the distribution remains the same. A higher potential will enlarge the local minimum, while a lower one will decrease it.

The field plate, being an equipotential plane, will flatten the potential distribution, especially when it is a wide plate. When thin oxide layers are employed, the effect of the field plate is more pronounced; the induction is stronger and the potential distribution beneath the field plate more flattened.

During the calculations it appeared that when, below full depletion, the bias voltage is quadrupled, the potential on the field plate has to be doubled, in order to keep the same form of the potential distribution, and therefore also of the electric field distribution (the actual magnitudes are then doubled). This square-root relationship, as will be explained in the next section, will change to a linear one way above full depletion. In both cases, the absence of surface charge is assumed.

When surface charge is also present, the separating field Ex is hardly affected by a change in the bias voltage, assuming that no electrons are captured in the local potential minimum beneath the field plate. However, as explained before, electrons are likely to be captured in that bucket and the bias dependence will therefore be regained, although possibly not as strong.

In the absence of surface charge, electrons will be captured beneath the field plate when a sufficiently high potential is applied to it. For negative potentials, holes will be captured and a channel between the strips formed (as in MOS-transistors). In the most exotic case of surface charge and a small field plate with a negative potential, it is even possible to get successively

80

81

free electrons, free holes and again free electrons along the surface between two strips (Fig.4.2.4f.).

The field plate can also be used to model the effect when the aluminum contacting of the strips is misaligned and covers the oxide layer next to the strips. No real influence could be observed (only at the surface between the strips could the effect be noticed as a smoother curvature of the electric field), so this misalignment is not a critical step in strip detector fabrication cycles.

4.2.4. Bias dependence

When the bias voltage was varied, it appeared that all potential, and thus also all electric-field distributions were, except for a scaling factor, identical to each other. No surface charge and the same geometry were assumed. In fact, the distributions were scaled proportional to the square root of the applied bias voltage. This will be discussed in the following. In the one-dimensional case, the potential V(y) in the depletion layer as a function of the depth y can be described as [4.21,4.23]

V(y) = *r (-1/* y* + Wy), (4.2.1)

W = ^ > b i a 8 > (4-2.2)

where q is the elementary charge, N the doping concentration, e the permittivity, Vhiaa the bias voltage and W the depletion layer thickness.

The potential distribution loses it two-dimensional character at about one gap length (Sec.4.2.1.). This means that in the area of interest y is smaller than, say, 15/mi. At a bias of 25 V the depletion layer thickness is already 180 pm (N = 1 x 10 cm"3, Eq.(4.2.2)). Eq.(4.2.1) can therefore be simplified to (y«W)

V(y) ~ MKy = ^ L y = Emax.y. (4.2.3)

The error which occurs is y/(2W)<4% (already in this 25 V-bias case). Eq.(4.2.3) shows that in this region the potential distribution will look as if it

Fig.4.2.4. The effect of a field plate. The oxide layer thickness was set to 0.2 nm. a)-d) are without and e)-h) are with surface charge (1 x 10 cm' ) . a) No field plate, the surface potential is 1.0 V. b) Field plate with a potential of 0 V. c) Field plate with a potential of 1 V. d) Field plate with a potential of 5 V. e) No field plate, the surface potential is 6.4 V. f) Field plate with a potential of 0 V. g) Field plate with a potential of 6.5 V. h) Field plate with a potential of 10 V.

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is a linearly increasing one, as the plots (Fig.4.2.1b., etc.) indeed show. Moreover, Eq.(4.2.3) (together with Eq.(4.2.2)) shows that the potential at the spot where the one-dimensional solution starts to hold increases with the square root of the applied bias voltage. This explains the dependence found.

At overdepletion Eq.(4.2.1) becomes

V(y) = Vtd(y) + Vhia*~Vid y, (4.2.4) Nth

where Wth is the detector thickness and fd indicates the situation at full depletion. (The Vtd(y) function is therefore like Eq.(4.2.1), with W replaced by Wth\'V{d and Wth are related to each other by Eq.(4.2.2).) Again, for small y this can be simplified to

V(y) * (£max,fd + Vh[ZVfd)y = ^ " J ^ y = £max^ (4-2-5> Wth "th

From Eq.(4.2.5) it can be seen that the square-root relationship starts to resemble a linear one, and becomes linear in the end, when Vbiaa»V{d. This was also confirmed by the calculations.

Because the one-dimensional solution is achieved at the depth of one gap length, it should also be possible to calculate the potential distribution by simulating the strip detector, as it is a device with a thickness of one gap length, which is biased with the voltage known from the one-dimensional solution at that depth in the original device. This has been verified and confirmed. The new device is now clearly overdepleted and this explains why a square-root relation is also found when a two-dimensional calculation is used (the square-root relation holds not only in the y, but also in the x-direction). The (Poisson) equation to be solved for the new device is

V V = =£-, V(Wth>neJ = Kbia8,new, (4.2.6)

where p is the space charge density; the potential of the strips throughout is 0 V. The potential is split into two terms, the potential just at full depletion and the remaining part:

V = Vtd + VR. (4.2.7)

This enables Eq.(4.2.6) to be split into two equations with different boundary conditions, one representing the (new) device at full depletion with space charge, and one representing the device biased by the remaining voltage ^bias.new- fd without space charge:

83

v V f d = -=£, VM(Wth>neJ = Vlt (4.2.8a)

V V R = O , VR(WthineJ = V2. (4.2.8b)

Vy is found by Eq.(4.2.2) (where W is the new wafer thickness), and V2 is the difference between the new bias voltage (found by Eq.(4.2.1)) and Vv Because V1«V2 and because the potential distributions vary monotonically, the form of the total distribution will be nearly set by Eq.(4.2.8b). The space charge has no influence on the distribution. Finally, because Khia„ n?w varies with the square root of Kbias, all distributions will vary with VKbias. It should be noted that this relationship only holds in the surface region, and not in the bulk, where the one-dimensional solution is valid. A low bulk doping concentration and a bias voltage which is not too low are also required for this relationship to be valid.

In the case of surface charge, the distributions are not influenced by the bias voltage. The separating field E^ is independent of the bias voltage. This is due to the fact that the field Ey at the surface is completely determined by the surface charge a [4.17]:

Ey(0) = f . (4.2.9)

Because the field in the y-direction at the height of the junction depth is much lower (although still varying with vVb i a 8), the gradient in this field is nearly independent of the bias voltage. This means that the Poisson equation becomes one-dimensional, and consequently, that the E^-field becomes bias independent:

?3L = -JLK. = - £ 5 L + JL = constant. (4.2.10) dx dx2 dy £

Because free electrons can compensate the surface charge effect, the distributions will probably still be influenced by the bias voltage.

4.2.5. Disturbance voltages on the strips

In operation, the strips of a strip detector have to be biased and connected to electronics circuitry for readout and further processing. Usually, the first stage in this electronics chain consists of amplifiers or multiplexers. The biasing is done either by bias resistors or via the input stage of the electronics. Due to differences in this input stage or in the bias resistors, the strip potentials will not be equal to each other. In addition, due to junctions being leaky or floating, disturbances of the strip potential can occur. They are limited by breakdown or forward biasing of the strip junction. In this

84

section the effect of these disturbances in the strip potential on the potential distribution are investigated.

As seen in the previous section, a square-root relationship was found between the potential distribution and the bias voltage below full depletion. This also holds for a disturbance voltage on a strip: To retain the same distribution when the potential of a strip is doubled, the bias has to be quadrupled. The explanation is the same as in the previous section. Way above full depletion the relation will become linear. When there is (positive) surface charge between the strips the Ex-field will remain unchanged for a fixed disturbance potential on one strip and a varying bias voltage, as the latter no longer influences the field (Sec.4.2.4.).

1.2 xicf

Fig.4.2.5. Only the collection volumes of the two neighboring strips are affected by a disturbance voltage (IV) on a strip. The bias voltage is 100 V.

Fig.4.2.6. The potential distributions at 25 V (upper three) and 100 V (lower three) bias voltage, when the middle strip has a disturbance voltage of 0, 0.5 and 1 V (from left to right). There is no surface charge.

oo

86

Fig.4.2.7flj FigA.2.7b)

0.6 1.2 1.8 2.14 0 0.6 1.2 1.8 2.U 3 3.6 4.2 4.6 5.U 6 6.6 7.2 1.6 «10*

Fig.4.2.8. Fig.4.2.7C,)

Fig.4.2.7. The effect of the gap width on the influence of a disturbance voltage (no surface charge). The bias voltage is 100 V; the disturbance voltage is 0.5 V. The pitch is 20 p.m for all three devices; the gap is a) 5, b) 10 and c) 15 \im.

Fig.4.2.8. Electric field distribution for a 40 \im pitch detector with a 5 \im gap; the other conditions are as in Fig.4.2.7.

87

Depending on the sign, the effect of a disturbance voltage on a strip will be the increment or decrement of the collection volume belonging to this strip and therefore the opposite change in collection volume of its two neighboring strips (cf. Fig.4.2.10.). It was found that the collection volume of the strips adjacent to these neighboring two are hardly affected (Fig.4.2.5.). This means that for a floating strip its charge is collected solely by its neighbors.

The border of the collection volume is defined by those places where the separating field E^ is zero. This border is aimed at being positioned midway between two strips and at being insensitive to external effects as much as possible. For this reason a high bias voltage is preferable, as is shown in Fig.4.2.6., where the potential distributions are plotted for 25 V and 100 V bias voltage. The potential of the middle strip is 0, 0.5 and 1 V, respectively. It can clearly be seen that this strip will collect less and less with increasing disturbance voltage. The effect is smaller at the higher bias voltage. Furthermore, the plots of a 0 V and 0.5 V disturbance at 25 V are clearly identical to respectively those of a 0 V and 1 V disturbance at 100 V bias, showing the square-root dependence.

The presence of (positive) surface charge also lowers the sensitivity to disturbances. Although the bias voltage no longer has any influence on the distribution, the shift in the £^=0-line is also less strong than without surface charge. In principle, the surface charge density can be enlarged by implanting extra donor atoms (n-type bulk assumed). However, the result will be the presence of free electrons at the surface (Sec.4.2.2.), and almost no improvement will be obtained (depending on the extent to which the free electrons are compensating the surface charge).

By increasing the gap width between the strips, the effect of a disturbance potential will be diminished, as shown in Fig.4.2.7. The small gap device will hardly collect any charge, whereas the large gap one is hardly affected by the disturbance. The field in the neighborhood of a gap is the same for devices with the same gap but with different pitches (Figs.4.2.7. and 4.2.8., Sec.4.2.1.). It is therefore possible for a disturbed strip in a large pitch device to collect charge, while it will not in one having a smaller pitch but the same gap. For at one gap length into the substrate the electric field is only in the y-direction and charge generated right in front of the wider strip will still be collected by that strip (Fig.4.2.8., the case of no surface charge is depicted). So, when this strip is a floating one, it will first collect that charge, inducing a (capacitive) pulse on its neighboring strips, after which the charge will flow to those strips.

4.2.6. Experiments

In this section the measurement results obtained on test structures are presented and discussed. They were fabricated at the IC-Workshop of the Delft University of Technology (Fig.4.2.9.). They were all of the junction-type, and made by process II or III (defined in Table 3.2.1.). High purity

88 1 2 - 3

n-type silicon with a doping of about 10 cm was used as bulk material. The structures consist of strip detectors having about 20-30 (p-type) strips with pitch sizes between 40 and 120 fim, while the gap in each category was varied between 10 /zm and that width, leaving a strip width of 10 /zm. The length of the strips was mostly 4.2 mm, but smaller lengths were also designed. The total size of the devices was such that they could be easily mounted in standard DIL-housings. Some detectors included field plates between the strips. Some diodes having the same area as the strip detectors were also designed. All devices had a guard ring surrounding the strips, in order to shield them from the effects of the surface and far bulk. In this section, only the measurement data from the four most instructive test devices will be presented. Their properties are listed in Table 4.2.1. (as the dopant atoms in process II also diffuse in the lateral direction, the gaps will be about 4 itm smaller than listed in the table). These devices all had the same area (about 6.7 mm2).

Fig.4.2.9. The fabricated wafers with the different detectors.

Table 4.2.1. Geometry data (in um) of those test devices whose measurement results are presented.

sample

A

B

E

F

pitch

50

50

100

100

gap

15

25

15

25

89

The first thing to be measured once a detector has been made is its /F-characteristic. For a strip detector this is accomplished by setting all strips to ground potential and applying the bias to the substrate. In this way the current through one as well as through several strips can easily be measured. The leakage current of the fabricated devices made in process II happened to be on the higher side: about 100 nA at 100 V was found for the total current of a device. In fact, these higher currents facilitated the measurements, as the strip's dc-impedance is lower (see below, Eq.(4.2.12)), which therefore puts smaller demands on the measurement apparatus. In this sense, the leakage current of the devices made in process III (about 1 nA at 100 V for the whole device) was often too low. Therefore, only the results obtained on the devices made in process II will be presented throughout this section.

Both the form of the /^-characteristics as well as the magnitude of the currents were the same for each design. The width of the gap had no measurable effect on the leakage current. (The presence of the Si-Si02 surface could be observed, although only as an increase in the leakage current when the field plates were negatively biased.) Conjointly, the /K-curves of the full-size diodes were identical to those obtained on the strip detectors. The current increases linearly with the area (larger strip length, larger pitch size, or more connected strips). Therefore, the one-dimensional theory of the leakage current, Sec.3.1., remains applicable to strip detectors.

The effect of the gap width on the electrical behavior of the detector was first investigated by measuring the dc-resistance of a strip. This resistance is defined as the ratio between a disturbance voltage AK, applied on the strip, and the current change A / due to this disturbance voltage through the strip, interpolated to the point AK = 0:

D def AF (4.2.11) AK=0

When the bias is sufficiently high, the AV-AI curve is fairly linear and i^,. is well defined.

The effect of a disturbance voltage on a strip is a change in the collection volume of that strip (see also Sec.4.2.5.). For a positive disturbance voltage, as depicted in Fig.4.2.10., this volume and consequently the leakage current will decrease. From Fig.4.2.10. and Eq.(4.2.11), this can be expressed as

where I0 is the leakage current through an undisturbed strip, p the pitch size and Ap the change in the pitch. This equation clearly expresses that for a

90

Uzzi X/T/A AV

Y/7/A

rAp rAp

TZ

Fig.4.2.10. The change in collection volume due to a (positive) disturbance voltage.

given leakage current a high dc-resistance corresponds to a low disturbance sensitivity and therefore a better position indication (the charge will be collected by the nearest strip).

This resistance is not an indication of the crosstalk effects between the strips (as it is not applicable for resistive charge division, Sec.5.1.1., either). Instead, that is an ac-resistance. This dc-resistance is due to a bulk effect. Only the part arising from surface resistance (channel) contributes to both kinds. On the other hand, however, the gap width is of influence with respect to the crosstalk, as, for example, the interstrip capacitance will be lower at larger gap widths.

Table 4.2.2. Eq.(4.2.13).

Measured dc-resistance (in Gtl) and constants fitting

sample

A

B

E

F

*dc

2.0

3.1

1.7

2.8

a c

0.74 0.31

0.64 0.86

0.68 0.38

0.62 1.04

In Table 4.2.2. the measured values of the dc-resistance are depicted for the samples made in process II. They were on the order of a few gigaohm. The leakage current of the devices made in process III was a hundred times

91

smaller. Accordingly, a hundred times larger dc-resistance was measured. The dc-resistance was inversely proportional to the length of the strips and, as shown by Table 4.2.2., increased with increasing gap width. It can be seen that the pitch size does not really influence this resistance, as (cf. Eq.(4.2.12)) the leakage current is proportional to the pitch size. Biasing the field plates did not show any effect on the dc-resistance value until the bias went below a certain negative voltage and a p-type channel was formed between the strips. The strips are short-circuited and a zero resistance is found. Varying the bias voltage had no influence on the measured dc-resistance of the devices made in process II. In theory, the change in collection volume Ap will be smaller at higher voltages (Sec.4.2.5.), and therefore this resistance will be larger. However, this holds only when the leakage current remains constant, as depicted by Eq.(4.2.12), and this was not the case with these devices. (The smaller change in collection volume at higher bias voltages could, however, be observed with the laser beam measurements, see below.) On the other hand, the leakage current of the devices made in process III saturated at a certain voltage, and consequently an increase in the dc-resistance was measured. Likewise, at the onset of breakdown the value of this resistance decreased rapidly. (In the case in which the strips are of the same doping type as the bulk material the influence of the bias voltage on the dc-resistance will clearly be more pronounced: below full depletion the strips are ohmically connected to each other by the undepleted bulk and a low dc-resistance is measured; above full depletion the situation will become similar to the usual case of strips of an opposite doping, and a strong increase in the dc-resistance will result [4.11,4.26].)

A second way to investigate the gap influence on the detector's electrical behavior is to measure the potential of a floating strip as a function of the applied bias voltage at the rear side (the other strips were set to ground potential). The impedance of the voltage meter measuring this potential was larger than 10 Ü. For the devices made in process II this impedance is large enough, as the dc-impedance of the strips is on the order of 10 n (Table 4.2.2.). However, for the devices made in process III this impedance was too low and no good measurements could be performed. (Physically speaking, it is required that the impedance of the voltage meter be so large that the current through the meter due to the potential rise of the floating strip is much smaller than the current I0 which the undisturbed strip can produce. By Eq.(4.2.12) it then follows that this impedance should be much larger than the strip impedance. When this is not the case, the strip will not be floating and there is still a compartment, Fig.4.2.10., in which the current is collected to supply the voltage meter.)

According to simple theory the floating strip potential will be a constant plus a term which is square-root proportional below and linearly proportional above full depletion with the applied bias [4.27]. In extreme cases, such as a very high surface charge density, the constant term may be dominating, and the potential of the floating strip will be bias independent.

92

* * * * * * GRAPHICS PLOT * * * * * * IS 423 Al

«KKWXJt GRAPHICS PLOT * * * * * * IS 483 Bi

.5000 /div

.0000 A

S \*s

1.000 /aiv

. 0000 / /

s"

****** GRAPHICS PLOT ****** IS 423 El

****** GRAPHICS PLOT ****** IS 423 Fl

/ / /

S

/div

.0000 / /

5.000/div [ V) VIN s.ooo/mv ( v) E F

Fig.4.2.11. The measured potential of a floating strip as a function of the applied bias voltage. Note the different scales.

The measured curves, four of which are shown in Fig.4.2.11., fulfilled the relation

V„ = cVt bias - (4.2.13)

The measured values for a and c are listed in Table 4.2.2. The maximum applied bias voltage was 100 V. Therefore, no data above full depletion were taken. According to the theory a has to be independent of the gap size (V2 below and 1 above full depletion), while c should increase quadratically with it [4.27]. The experiments showed that a was fairly constant, while c clearly increased with increasing gap size. (This occurred exactly quadratically when observing devices with .the same pitch.) From the measurements it can be concluded that the interaction between the strips is less at larger gap widths. It can also be seen that the pitch size is not really important with respect to this interaction.

93

No effect was noticed in biasing the field plates, except when this bias became negative. Then, the potential of the floating strip increased less with the applied substrate bias and it ultimately even stayed at 0 V. This again demonstrates the formation of a p-type channel between the strips. When the complete channel is ultimately formed, the strips are short-circuited and the strip under measurement, no longer floating, is kept at 0 V.

Next, capacitance measurements were done. Firstly, the capacitance of all strips in parallel as a function of the bias voltage was measured (Fig.4.2.12.; stray capacitances have been accounted for in this and the subsequent plots). This showed an anomalous effect, which was difficult to comprehend. The capacitance decreased less than would be expected from one-dimensional theory, but around 14 V the capacitance decreased faster and regained its

0. I i I I j I I 0 . 10. 2 0 . 3 0 .

bias voltage [V]

Fig.4.2.12. Typical CV-curves for the strip detectors (with the strips in parallel, curve ABEF) and for the full-size diode (curve D).

94

predicted value (Fig.4.2.12.). The full-size diodes, having the same area as the strip detectors, did not show this effect at all (curve D, Fig.4.2.12.). Consequently, care has to be taken when using a microstrip detector itself to characterize its own diodes (e.g. in lifetime measurements, Sec.3.1.3.2.).

Clearly, the dip is due to the strip pattern. It is believed that it is due to the presence of free electrons between the strips (Sec.4.2.2.), which are acting analogously to the effect of surface states when measuring MOS-capacitors [4.21,4.23,4.28]. As explained in Sec.4.2.2., these electrons are due to surface charge, which creates a local minimum in the potential distribution. The height of the potential minimum is fixed by the amount of surface charge, while its content is dependent on this height and the applied bias voltage (cf. Sec.4.2.4.). When the phase of the measurement signal is such that it increases the total bias voltage, the content of the local minimum will decrease and some electrons will no longer be captured at the surface. They will drift to

0. 10 . 2 0 . 3 0 . 4 0 . 5 0 .

bias voltage tV]

Fig.4.2.13. CV-curve for one single strip.

95

2 . 2

2 . 0

CU O c co ■i-3 £ i 2 . 2 U CD Q . CD O

2 . 2

_ 2 . 0

0 . 0 10 . 0 2 0 . 0 3 0 . 0 0 . 0 1 0 . 0 2 0 . 0 3 0 . 0

2 . 0

1 . 8

1 . 6

1 . 4

1 . 2

1 1 ' 1 ' 1

: l l lui, ' ■

; ' ;

h 1 1 1 1 1 1 0. 0 1 0 . 0 20. 0 30 . 0 0 . 0 1 0 . 0 2 0 . 0 30 . 0

bias voltage [V]

Fig.4.2.14. The "interstrip capacitance".

the n-contact and compensate the bias increase by the measurement signal. Consequently, a larger capacitance will be measured. At 14 V bias voltage the content of the local minimum has decreased to zero and the capacitance value of a plane diode is measured. Likewise, the dip disappeared when the field plate was negatively biased. When the measurement signal is in the other phase, the minimum is filled again by thermal generation. The dip disappeared when the amplitude of the measurement signal (normally 0.1 V) was increased, because the thermal generation then became too small to compensate the first half phase. It was impossible to observe the same effect by increasing the measurement frequency, as it (100 kHz) could not be increased further.

Secondly, the capacitance of one strip was measured, with the other strips connected to ground potential (the bias was applied at the n-contact). As is expected, the samples with the larger pitch (i.e. larger area) have the larger capacitance, while for those with the same pitch the smaller gap (larger

96

interstrip capacitance) results in a larger capacitance (Fig.4.2.13.). These CF-curves showed an even more pronounced step at 14 V (Fig.4.2.13.).

By positively biasing the field plates the conductance in parallel with the capacitance increased, whereas in the other case it decreased until a channel was finally formed, resulting in a strong increase in the conductance. The measurement system could not measure the conductance in parallel with the capacitance with a high accuracy, but it could be seen that this conductance decreased with increasing bias voltage. (Light of course increased the conductance.) These results are also in accordance with the hypothesis of free electrons being present at the surface.

It is assumed that the measured strip capacitance is the combination of the interstrip capacitance and the (one-dimensional) junction capacitance in parallel. In this way the interstrip capacitance was calculated from the two measurements (the junction capacitance is assumed to be proportional to the number of strips). As was to be expected, a fairly constant value for this capacitance was found, except, again, for a step at 14 V bias (Fig.4.2.14.; the capacitance values are on the order predicted by theory [4.18,4.29]). It can be seen that this interstrip capacitance is the same for the devices with the same gap width, being smaller for the larger widths. It can also be seen that this capacitance is much larger than the junction capacitance (except at low bias voltages). These conclusions are in agreement with expectations.

Finally, the detectors were tested with three kinds of radiation; experiments were carried out at CERN with 5 MeV a-particles and minimum ionizing particles, and at Delft with laser light. With the minimum ionizing particles and the laser light the position resolution of the different detectors was examined. These tests show that the position precision is solely determined by the pitch. It is not affected by the gap width. However, the existence of some gap width (i.e. some resistance between the contacts) is essential. A plane-diode with strip-like contacts has no position resolution.

The border between two collection regions is in the middle of the gap. The measurements with the laser beam showed, in accordance with the calculations (Sec.4.2.5.), that this border moves when a disturbance voltage is applied to a strip. This movement is smaller at larger gap widths, or at higher bias voltages. Only a movement of the borders belonging to the disturbed strip could be measured; the others remained unchanged.

While the position precision is independent of the strip width, the a-spectra obtained (cf. Fig.2.2.) were not, see Fig.4.2.15. A second peak arose to the left of the main peak (lower energies). This peak moved even more to the left when the gap increased. It also became wider. The main peak remained on the same spot with the same FWHM, as for the full-size diode. When the pitch was doubled at constant gap width, the height of this second peak was halved. The results presented in Fig.4.2.15. are obtained by connecting all strips in parallel. However, in the case of the measurement on one single strip the results were the same. The spectrum obtained on the full-size diodes did not show the second peak.

97

Fig.4.2.15. Spectra obtained with 5 MeV a-radiation.

This phenomenon is explained as follows. It is assumed that due to surface charge there are free electrons between the strips (Sec.4.2.2.). When an Q-particle impinges the detector on a strip, nothing special happens. There is electron-hole pair generation. The electrons drift into the substrate, and the holes are collected by the strip. However, when the a-particle impinges between the strips, the holes can recombine with the free electrons at the surface, during their movement to a strip. This results in a lower output pulse and, consequently, in a count in a lower channel number. When the gap between the strips is larger, there is more room for such recombinations and the second peak will move to lower energies and become wider. When the pitch is doubled, the number of these recombinations is halved and the second peak will be half as high.

4.3. CONCLUSIONS

In this chapter the pulse responses on a strip detector's strips and the influence of the strip width on its performance have been analyzed. As such a detector is intended to be used for position determination first a short discussion of the position precision of the device was given.

98

The shapes of the pulses have been calculated by a novel computation method [4.1]. Because it is based upon Tellegen's theorem [4.2], this method is more general in character than those proposed earlier [4.15,4.16], making it applicable to other induction problems as well. The induction at a certain strip can be calculated by scaling all currents flowing inside the detector (resulting from the movement of the generated charges) by a reference field and then integrating these weighted currents. This field depends solely on the geometry of the detector; bias conditions etc. are of no importance. It can therefore be determined separately, after which it can be used for every conceivable pattern of currents and generation sources.

Three main conclusions could be drawn from the computed pulses. First, they showed, as was anticipated, that for the most part during the collection of generated charge the current induced in a strip next to the one going to collect the charge is the same as the "real" current from that collection strip. Secondly, it appeared that the induction on the neighboring strip is the strongest when the charge has been generated at the surface of the detector. When the charge is generated throughout the whole device (line-like source), this induction is the weakest. And thirdly, it became clear that the compensating ("negative") part of the pulse, induced in the strip next to the one going to collect the generated charge, will become a short, sharp dip in the case where the junction is on the side of the strips and the charge is generated at a single (point-like) spot.

The effect of different geometries on the performance of a strip detector had been investigated, both theoretically and experimentally. The results were in agreement with each other. It was found that the pitch determines the position resolution. With respect to the electrical behavior, the gap is the determining parameter, rather than the pitch. A strip detector can better be seen as a single plane detector with disturbances formed by the gaps, rather than as a collection of strip-like detectors put in a row. Only when the influence of disturbance voltages on the strips is considered is the strip width of importance with respect to the position indication of the detector. A wide gap diminishes this influence, and therefore such detectors will have a smoother position distribution. A higher bias voltage also reduces this disturbance influence, but can, in the case of less ideal devices, thereby also increase the leakage current. Moreover, the collection time will then be shortened, thereby reducing the outdif fusion of the carriers, which could be necessary to achieve a better position estimate [4.5].

In fact, the gap determines the potential and electric field distribution in the neighborhood of the strip, independently of the pitch size. About one gap length into the substrate, the distributions already equal those of one large plane (one-dimensional) diode. Therefore, it is possible that charge generated in the bulk can diffuse out during its drift towards the strips in such a way that it will be collected by more than one strip.

A large gap seems to be preferable. The interstrip capacitance will be lower, thereby reducing the total strip capacitance, which is important for noise considerations (Sec.2.2.3.). A lower interstrip capacitance also means a

99

lower crosstalk. Moreover, the influence of disturbance voltages will be less. The largest gap obtainable, given the desired pitch (which means the smallest strip width obtainable), is determined by the technology used (masking, alignment, minimum contact windows, and so on). Some drawbacks of a large gap can be an increased appearance of surface effects, such as surface generation of leakage current (which was, however, not measured), and trapping or recombination [4.21,4.23]. Besides, the collection time can be larger due to the longer path over which the carriers have to move. Furthermore, the application of capacitive charge division, requiring a specific interstrip capacitance value, Sec.5.1.1. [4.30], may influence the design of the gap width.

The existence of surface charge did not seem to have any effect on the performance of the detector, nor did the field plates. In principle, the influence of the bias voltage and of disturbances in the strip potential will be less with (positive) surface charge (or with a positive field plate potential). However, free electrons will be attracted to the surface and will compensate the surface charge (or field plate potential), thereby reducing the effect of the surface charge (or field plate). Only when the radiation to be detected generates a considerable amount of electron-hole pairs in the neighborhood of the surface can the presence of surface charge (field plates) be noticed.

100

CHAPTER 5 READOUT

5.0. INTRODUCTION

This chapter will deal briefly with the readout possibilities of a microstrip detector. When the output pulses of all strips are read out and processed the highest position resolution is obtained. This implies, however, that a lot of electronics circuitry is required, and moreover involves the problems of how.to connect all the strips in a reasonable way and how to place all that circuitry in the space available (not to mention the costs). As far as the second point is concerned, research is being directed towards monolithic integration of fast, low-noise preamplifiers [5.1,5.2]. As for solving the first one the possibilities of on-wafer integration [5.3,5.4] (Sec.3.2.5.), or the use of elevated connecting schemes [5.5,5.6] are being studied.

Clearly, some kind of multiplexing scheme has to be used, in order to reduce the number of output nodes. Several schemes have been invented (see Sec.5.1.), but which one has to be used depends on the kind of experiment in which the detector is to be implemented. (The use of detectors like the silicon drift chamber or CCD detector also has to be considered, Sec.2.3.4. [5.7,5.8,5.9].) For example, the so-called fixed-target and collider experiments are known in high-energy physics (see Chapter 1), each of which places its own constraints on the detector set-up [5.9]. The event rate and multiplicity are quite different in the two experiments, allowing room for more or for less multiplexing. Further, the space available for electronics and/or cabling is completely different in both types of experiment, as are the requirements put on the system by multiple scattering and radiation damage problems.

In the next section, Sec.5.1., some schemes for reading out the strip detector will be reviewed. In the subsequent section, Sec.5.2., the use of CCDs for the readout will be considered separately.

5.1. READOUT SCHEMES

5.1.0. Introduction

Up to now, a number of readout schemes have been invented. As they have been discussed extensively in various overview articles [5.7,5.8,5.9,5.10], they will only be summarized in this section.

The different schemes can be divided into two groups: the passive and the active kind of readout. The first only makes use of passive resistor-capacitor networks and can therefore be implemented in the detector design without a great deal of effort. It thus also solves the interconnection problem. The

102

second makes use of electronics circuitry for the multiplexing. In such a scheme the input-output function can be controlled much better, but, due to the increased processing complexity, the integration of the detector and the multiplexing electronics on one wafer is far more difficult. The next subsection will deal with the passive multiplex schemes first, after which the active ones will be brought up for discussion.

5.1.1. The passive readouts

The methods based upon charge division constitute the main representatives of the group of passive readouts. They are based upon the principle of reading out only every « t h strip. Charge collected by the intermediate strips is read out by the two neighboring readout strips. The way in which this charge is divided over those two strips indicates the position of the collection strip. (In order to avoid crosstalk with other readout strips, the preamplifier should have a low input impedance compared to the interstrip impedance, as it is assumed to be in the following.) Depending on the value of the combined interstrip /?C-product with respect to the amplifier shaping time constant the charge division is either capacitive (larger ^C-value) or resistive (larger shaping time constant). The interstrip RC-time is composed of the product of the total interstrip ac-resistance and capacitance (Sec.4.2.6.) between the readout and collection strip.

Resistive charge division, the first multiplex method to have been employed, is frequently used in nuclear physics applications of solid-state detectors [5.11,5.12]. Closely related to this method is one based upon the charge diffusion time [5.13]. In both methods the strips are connected to a resistor line (e.g. implanted conjointly with the strip layer itself), which is read out on both sides. In the first this line is terminated by a low impedance and the position is determined from the ratio of the charge flowing out on both sides, while in the second the line is terminated into its characteristic impedance and the time delay between the arrival of the output pulses on both sides depicts the position. (Due to the distributed strip capacitances, the resistor line represents a diffusive RC-Yme [5.13].)

There are some practical advantages in each of the methods [5.14]: The diffusion time method requires uniformity and stability of both the electrode resistance (/?D) and capacitance (CD), as these determine the .RC-delay constant. In the charge division method the position determination is independent of (linear) variations in the total electrode resistance (such as with time, temperature, and so on). Also, a nonuniform distribution of the electrode capacitances is allowed. The resolving time for the charge division method is about one electrode time constant -RDCD, for optimum position resolution and linearity. The resolving time for the diffusion time method is several times this time constant #DCD. The charge division method is inherently linear with optimum filtering. The diffusion time method is nearly linear, provided the diffusive line is terminated into its characteristic impedance and preamplifiers with . a high input impedance (and short

103

connections to the detector) are used. The charge division method requires preamplifiers with a low input impedance, and they can be some distance away from the detector (limited by pickup noise from the surrounding electrical equipment). The diffusion time method is not suitable for detectors with a long and variable charge collection time, which impairs the timing accuracy, while the charge division is independent of the charge collection time. The charge division requires a divider, either in hardware or software, while the diffusion time method requires a time digitizer. This last point is sometimes decisive when having to choose between these two methods [5.14].

Both methods have found widespread use in making position-sensitive detectors for nuclear physics [5.11]. In the case of microstrip detectors it is also possible to use the method of capacitive charge division [5.15,5.16]. The obtainable position resolution is about the same as for the foregoing two readout schemes [5.13,5.17]. Which one is to be used is a question of experiment design.

Capacitive charge division is possible, because the capacitance between the strips is now much larger than that of the strip to ground, see Sec.4.2.6. This means that the charge collected by a floating strip will be nearly completely induced in the readout strips. By enlarging the interstrip capacitance the signal attenuation due to the charge sharing between this capacitance and the strip capacitance to ground will be further diminished (and thus the linearity will be improved). Such an increased capacitance, on the other hand, will also increase the noise (Sec.2.2.3.), which will result in a decrease in the signal-to-noise ratio. Therefore, an optimum interstrip capacitance value can be found, resulting in the highest position resolution [5.16].

The admittance of the interstrip capacitance can also be increased by a parallel resistor. To be active, this resistor should be smaller than the impedance of that capacitance (i.e. their i?C-time should be smaller than the amplifier shaping time constant); the charge division has simultaneously been turned over from capacitive to resistive. (To be really effective for resistive charge division, the RC-time of this resistance and the strip-to-ground capacitance - the above-mentioned Rj)CD response time - also has to be smaller than this shaping time constant. In microstrip detectors, however, this condition is automatically fulfilled when the first is.)

The effect of charge division will be a reduction in the number of output nodes, yet a decreased ability to separate closely spaced tracks. The multiplex scheme to be discussed next, on the contrary, requires that particles hitting the detector simultaneously have to be within a region no wider than a "band" (defined below), in order to be able to determine their positions of incidence. The basic concept, Fig.5.1., is to divide the strips into groups. The first strips of each group are connected to each other, as are the second, the third, and so on. A second readout has to be made, each channel of which, labeled a "band", covers a group of strips. When a particle hits a strip, not only that strip, but also its corresponding band, will give an output pulse. The position of the particle can then be deduced from the combination of the strip and

104

1— = t " I l ~ 1 1 ____ll____ll_-_-ll_._-ll S T R I P S

BANDS Fig.5.1. The basic multiplex concept.

band number. In this way, N strips can be read with a minimum of 2VN connections.

The second readout can be made either directly or capacitively. When the primary strip groups are made as separate detectors, the direct readout can be accomplished by mounting these detectors together, reading out each rear side as a separate band [5.18]. Clearly, for microstrip detectors this method is not really advisable (one can try to make the bands by masking the rear side during processing).

A better solution is found in capacitive readout. The use of a second metalization layer allows this readout, based upon the coupling between this and the first metalization layer, to be realized [5.19]. The strips, connected as before, are made in the first interconnection layer, while the bands are made in the second, see Fig.5.2. Depending on the process used, this layer will lie 1 to 4 /zm above the strips. The capacitive coupling between the bands and the strips is responsible for a band's being able to output a pulse when a strip below it is doing so. (The strips of the different groups belonging to each other have to be connected by series resistors to ensure a coupling of each strip to only one band - the readout amplifiers are of course assumed to have a low input impedance.) Because the method uses standard IC-technology processing steps, it can easily be included in the strip detector fabrication process.

Fig.5.2. The multiplex principle capacitively applied by the use of a second metalization layer.

105

The concept has been tested [5.19]. First, the device was simulated by the program SPICE. Each strip was represented as a capacitor. Clearly, resistors designated their series resistors, as capacitors were the coupling between the strips and bands. In the program, a capacitor was also placed between neighboring strips, simulating their mutual interaction. The outputs were connected via a small resistor to earth, representing the input impedance of the preamplifiers. A current pulse was applied on one strip node and the pulses on the associated and neighboring outputs were computed. The results confirmed the principle: upon the right choice of the series resistance and coupling capacitance, large pulses at the former and small ones at the latter outputs could be achieved.

:. . 4 f : " ■ ■ ■ ■ '

Fig.5.3. Photomicrograph of the chip.

Next, a test device was fabricated at Philips Industries in Nijmegen. It consisted of 49 strips, divided into 7 groups of 7 strips, thus having 14 output nodes (Fig.5.3.). The p-type strips, made in an 8 /im thick n-type epilayer, were 2 mm long and 10 pm wide at a 20 jum pitch. The series resistors and coupling capacitances were designed according to the optimum values found by the SPICE simulations.

Because the device had only an 8 /an deep detection volume, it had to be tested with a-particles (the epilayer was necessary for the other structures on the wafer). This test was carried out at CERN with a Am a-source, which irradiated the detector uniformly. All 14 nodes were read out. The signals from one strip or one band were used as a trigger. In this way, it could be

■ ■ • i n

u , i i ■■ '■! * " ■ i.i'Z. ■-H J i '■ i ' i ' ■i ;-n H m i U I . . I I I-

* ,"IIL "* ■

: . ■ . : " ■ ' ■ ■■

*

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seen that per event only one strip-band pair was hit (Fig.5.4.), in confirmation of the concept. The crosstalk between neighboring bands was less than 4%. For the strips, however, this was 20%, which is possibly due to the occurrence of double hits (cf. [5.20], where a similar distribution was found), which, of course, cannot be distinguished from true crosstalk signals (according to the SPICE calculations, the crosstalk enhanced by the series resistors between the strips and amplifiers should be excluded).

1007.

507

07

a)

1007 -i

507

12 3 4 5 6 7 STRIPS

2 3 1 5 6 7 BANDS

07

b) 1 2 3 4 5 6 7

STRIPS 1 2 3 4 5 6 7

BANDS

Fig.5.4. The number of pulse counts on the strips and bands a) when triggering on strip #5. b) when triggering on band #5. It appeared that strip #2 was not well connected.

The capacitive coupling can also be used for other applications. For example, to have a trigger to determine whether a particle has hit some strip of the strip detector, a "band" which covers all strips can be used. Another application is in two-dimensional position measurement, [5.19,5.21].

5.1.2. The active readouts

The active readouts, making use of electronic circuitry, constitute the more sophisticated readout schemes. Experiments have been conducted with a so-called track-and-hold circuit [5.22] and with the use of shift registers, where the information is charged in parallel and extracted in series [5.23]. Similarly, the feasibility of using CCDs to serialize the microstrip information has been studied [5.24] (see also Sec.5.2.). Finally, full electronic sample-and-hold multiplexers have been designed (in NMOS and CMOS) and tested [5.5,5.20,5.25].

Due to differences in their processing, integrating the detector and the readout on one wafer constitutes a technological problem. As a result, until

107

now both parts have been made separately, leaving the problem of connecting all the strips unsolved. To this end, preliminary studies have been performed to investigate the possibilities of on-wafer integration, Sec.3.2.5. [5.3,5.4], or the possibilities of advanced interconnection schemes between the detector wafer and the electronics chip [5.5,5.6].

For complete compatibility, however, not only should the processing be congruous, but the physical aspects should also fit. For example, a p-type strip will collect holes, which results in an output current in the outward direction of the strip. The input of the electronics should correspond to this current direction. A more serious point concerns the bias supply: A bias on the order of 100 V is required for the detector part, whereas for the electronics part it will not exceed the level of 1 5 V [5.4]. Furthermore, the bulk doping concentration is different in both parts. HP-Si is needed for the detector part (Chapter 1), whereas for the electronics a higher doping concentration is required. (This latter is a trade-off between device density, which is limited by the punchthrough of the depletion layer between neighboring diffusion areas, and the junction capacitance, which is important for the switching speed and circuit bandwidth [5.4,5.26].)

With respect to the processing compatibility, Sec.3.2.5. [5.3] seems to indicate that the entire electronics circuitry can be implemented along with the detector on the same wafer, as long as after the implantation of the detector junctions no treatments above 600 °C are applied to the wafer. Thus, diffusions, e.g. n- or p-wells, and implantations to be annealed at, say, 900 °C must precede the detector-junction implantation.

The biasing problem can be overcome by using some kind of isolation technique. Two principle methods can be designated: junction and dielectric isolation [5.4]. Junction isolation is based upon the (deep) diffusion of a p-well (assuming n-type detector silicon), in which the electronic circuitry is placed. In this way the "bulk" doping for the electronics is enlarged automatically. A drawback is the demand on the well-substrate junction, for it has to block the reverse bias of the detector (the strips and the p-well are at ground potential; the electric field is the largest at the junction). The applicability of this method has been demonstrated for the case of a deep-depletion CCD with the readout electronics made in a well of higher doping [5.26].

Dielectric isolation is based upon the (deep) implantation and subsequent annealing of nitrogen or oxygen atoms, resulting in a buried isolation layer [5.27]. In this way the biasing of the detector cannot affect the electronics part anymore.

A solution to the readout problem concerns the use of CCDs as parallel-in serial-out multiplexers, as they are used in the CCD detector itself. In comparing the strip and the CCD detector, see Sec.2.3.4., it appears that the strip detector is a fast device, but one having a lot of output channels, whereas the CCD detector, requiring only a few connections, is a relatively slow device. By combining these devices the readout time and number of output nodes can be optimized, see Fig.5.5. [5.28,5.29]. Additionally, CCDs

108

are attractive because they can be used not only for multiplex, but also for delay and memory purposes, so that only those data of interest, after receiving a trigger signal, have to be outputted, and those of no interest can be dumped on chip without being read out.

Fig.5.5. Schematic drawings of a) the strip detector, b) the CCD detector and c) the combined strip detector with CCD readout.

When fabricating such a combined device, two requirements must be fulfilled (in addition to those like noise and dissipation control [5.24,5.30,5.31]): all the signal charge collected by a strip has to be drained into the CCD; and this has to happen in one or a few clock periods. These two aspects will be discussed in the next section.

5.2. CCD READOUT

5.2.0. Introduction

Usually, a. Charge-Coupled Device is made up of an array of MOS capacitors [5.31,5.32,5.33], but pn-junctions can also be used to construct a CCD, which is then referred to as a Junction CCD (JCCD, [5.34,5.35]). In Fig.5.6. a cross section of a JCCD is given. In normal CCD operation both the

109

epilayer-gate as well as the epilayer-substrate junction are reverse biased. By applying mutually phase-shifted clock pulses to the p-type gates, traveling potential wells are induced in the epilayer, in which charge (electrons) can be transferred. The highly doped n-region, Fig.5.6., is a source of electrons. When it is on the proper potential, charge can be brought into the CCD by pulsing the first gate [5.32,5.33].

source o

gate o

gate o

drain o

\^_J \^__J

n-epilayer

p-substrate "

Fig.5.6. Cross section of a JCCD with a conventional input.

To have more control over the amount of charge that will flow into the CCD, a lot of input schemes have been invented [5.30,5.32,5.33]. All these methods suppose that the source is at a known fixed potential (the CCD input can be modeled as a high impedance; strictly speaking, when modeling the charge transfer, the current and continuity equations have to be solved [5.32]). A detector's output behaves, however, as a current source with a (relatively large) parallel capacitance, so that clearly some impedance matching is required [5.30] (i.e. the CCD input has to be made current instead of voltage sensitive [5.24]). Direct coupling is possible in the case of an integrated (photo-) detector array [5.30], but not very useful for strip detectors, as they collect a small amount of charge on a relatively large strip capacitance. Consequently, only a small rise in the strip potential, and thus the CCD input voltage, will occur. Moreover, due to the relatively large strip capacitance and high CCD input impedance a long input time will result [5.24]. So, in this way the generated charge collected by the detector cannot be brought into the CCD within a few clock periods and the signal will be lost.

Clearly, some kind of active input structure has to be used, one which of course requires no more area, as is determined by the strip pitch of the detector. Amplifiers, fitting in this space, have been designed, which enlarge the detector output signal and match the impedances of the detector and CCD [5.24].

An interesting solution is found in the use of the injection properties of the np-junction [5.35]. In the first gate of the JCCD an n-diffusion (or implantation) is made, configuring the source, see Fig.5.7. This results in an npn-transistor, whose collector part is formed by the CCD channel. By

110

forward biasing the np- (emitter-base) junction, charge will be injected into that channel independently of the CCD settings themselves [5.35]. Clearly, this input structure, called an injector [5.35], does not take up much wafer area. By adding p-junction formation to the fabrication process, it can be implemented in the case of an (n-channel) MOS-CCD as well.

emitter <?

\

base p

gate gate drain

XnTj-I L

MP M- 7

n-epilayer

p-substrate

Fig.5.7. Cross section of a JCCD with an injector input.

The input transistor operates in Common-Base mode. (Consequently, the input node is no longer floating, but is set by the base potential.) For proper operation it should be biased, as otherwise the signal charge would be lost in switching this transistor on. Further, the small-signal input impedance re of such a transistor is inversely proportional to its bias current / E (the other symbols have their usual meaning):

kT "e Qh'

(5.1)

The generated charge collected on the detector's capacitance CD will therefore flow with a time constant reCD into the CCD channel. This time constant should be smaller than the clock period in order to get all the charge together within such a period (see the next section). Therefore, a high bias current is preferred. The detector leakage current is not controllable in the sense that it can serve as bias supply (besides the fact that it is much too small to deliver this current anyway), and an external bias is required. This can be accomplished with the aid of current mirrors (enabling all inputs to be biased at once).

When an n-channel CCD is used together with a detector of p-type strips, Fig.5.8., the signs of the current flows are opposite, which again requires an external bias. In such a scheme, the current source is continuously injecting charge packets into the CCD, whose content is lowered when charge is generated in the detector. This has the additional benefit that the bias current can be set to the allowable maximum. This maximum is determined by the charge handling capability and gate area of the CCD, as will be seen in the next subsection, which deals with the injection time for this

I l l

configuration [5.36]. In the subsection thereafter, the coupling principle will be demonstrated experimentally [5.28,5.29,5.36].

V H 7

■4>i

<f>2 drain

v ^ p j v pjv pj i p; v ; ; ; ^ X$ET

n - epilayer p - substrate

a) bias

e T\Z

b) Fig.5.8. a) The detector-CCD coupling in principle, b) The equivalent circuit.

5.2.1. Injection time

Fig.5.8b. depicts the input scheme. The (differential) resistance re of the transistor is given by Eq.(5.1). When the voltage swing on the detector-injector node due to the charge collection in the detector is high enough, the injector will be switched off; instead of discharging through the resistance, the capacitance is then discharged by the biasing current source. At t=0 the capacitance is charged. If the transistor is switched off, the discharge time is

' o = G CD

= V(0) 'bias 'bias

(5.2)

where Q is the generated charge (positive number) and V(0) the voltage swing on the capacitance node at /=0. The discharge time is now defined as that time needed for discharging the capacitance for 98%. This time, when the transistor remains switched on, is then given by

112

kT Q> 'bias

= 4reCD = 4^-f^. (5.3)

To give a formula incorporating both switch states, the transition from one state into the other is assumed to be abrupt and to occur when the voltage swing on the capacitance node is, arbitrarily, equal to kT/q. (In fact, the transition will be gradual, as is accounted for in the SPICE simulations below; a diminishing part of the current from the current source will decharge the capacitance directly, while the other, growing part will bias the transistor, thereby similarly lowering the resistance re.) With the definition given above, the discharge time becomes:

kT Q) t2 = (3 + H- \nn)^-j^-, (5.4)

'bias in which n> 1 and is defined by

Q = n^-Qy. (5.5)

For large n Eq.(5.2) results, while for n=\ Eq.(5.3). The generated charge will be spread over either t0/T, tx/T, or t2/T clock

periods (depending on which of the Eqs.(5.2)-(5.4) holds; T is the length of time of one clock period). To determine whether it will flow into the CCD within one clock period (a ratio smaller than 1), the current /bias in the above formulas has to be replaced by the amount of injected charge Qmi within such a period T:

öi„j = w r . (5.6)

This amount is limited by the charge handling capability Qs and the gate area A of the CCD:

Öm a x = QeA. (5.7)

By substituting /bias in Eqs.(5.2) and (5.3) according to Eq.(5.6) it can be seen that injection within one period occurs for

kT Q < Öinj, 4 ^ " C D < Öi„j. (5.8)

Bearing in mind that the signal charge is subtracted from the injector charge, the first of these two is clearly a logical restriction. It shows (together

113

a)

125 MV STEP

b)

Fig.5.9. a) Response to a 2.5 mV step (lOfC). Horizontal: 0.2 ps/div., vertical: 0.5 mV/div. and 5 nA/div. b) Response to a 125 mV step (0.5 pC). Horizontal: 0.25 p,s/div., vertical: 20 mV/div. and 50 nA/div.

114

with Eq.(5.7)) that the minimum allowable gate area depends on the detector output signal and the CCD charge handling capability Qs. The second restricts the total detector capacitance, including wiring capacitance, and the CCD gate area to be designed according to this equation, given the charge handling capability of the CCD.

According to this simple theory, at low generation levels {Q<kT/q CD), the injection time is constant (Eq.(5.3)), whereas at higher levels it increases (Eq.(5.2)). In other words, the injection time is roughly proportional to the amount of generated charge with a lower limit given by the RC-time tv

The theory was confirmed by the simulation program SPICE. The capacitance CD, Fig.5.8b., was set to 4 pF and the current source /b i a 3 to 0.4 /J.A. The results are shown in Fig.5.9., corresponding to Eqs.(5.3) (charge injection of 10 fC) and (5.4) (charge injection of 0.5 pC, n=5), respectively. Note that in the latter the transistor is indeed switched off (there is no current through it).

In the case of negative output pulses (n-type strips instead of p-type) only discharge according to Eq.(5.3) can occur. At high output voltages the input transistor will then be switched to a higher current, thereby decreasing the resistance re (Eq.(5.1)). Because of this decrease injection will take place in a much shorter time than in the case of positive pulses. However, the generated charge will be added to the charge injected at equilibrium instead of subtracted from it. Therefore, the signal can be lost by overfilling the CCD. This was again confirmed by the SPICE analysis. So, depending on the maximum signal to be measured the bias current has to be set. Larger signals require lower bias currents. However, at lower currents the transistor characteristics are affected, deteriorating the injection properties. In the extreme case of zero bias current all or a substantial part of the generated charge will be lost' in switching the injector on.

5.2.2. Experimental results

The coupling principle has been verified by experiment, by wiring a separate detector to a JCCD [5.28,5.29] as well as by integrating the detector and the JCCD on one single chip [5.36].

In the first, the detector was irradiated by a fast LED simulating particle radiation. The detector bias was 30 V. At this voltage the detector capacitance was negligible compared to the large wiring capacitance of about 14 pF. The JCCD was run at 4 MHz, in three-phase operation. Fig.5.10. shows the CCD output signal after a 15 ns pulse of the LED, generating a charge of about 15fC in the detector (resulting in a voltage rise of about 1 mV, which therefore means that Eq.(5.3) is selected to express the injection time). The bias current was 1.5 /^A, yielding an input impedance of about 17 kfl. So, the time constant reCj), Eq.(5.3), was about 250 ns. This was also measured, as can be seen by inspecting Fig.5.10a. (the time between two dips is 250 ns). The impedance of the current source /bia8 is assumed to be large compared to re, and therefore without influence on the signal flow. (In the experimental

115

a)

b)

c)

Fig.5.10. CCD output signal (upper trace) and LED input voltage (lower trace) in the case of a separate detector and CCD. a) lbias= 1.5 IJ,A, V b i a g = 5 0 F , b) I b i „ = 0 .5M. V b i a s = 5 0 F and c) I b i a 8 -7 .5 /^4 . Vb i a g = 0 V. Horizontal: 1 us/div., vertical: 20 mV/div.

116

set-up a 5 Mfi resistor was used to realize the current source.) As the current gain of the injection transistor is well over 50, almost all charge will flow into the CCD-channel.

The influence of the bias current 7bias is illustrated in Fig.5.10b. The bias current is 0.5 /zA in this case, and as can be seen by inspecting the figure, the injection time has almost tripled (cf. Eq.(5.3)). Fig.5.10c. shows the influence of a larger detector capacitance. When the detector was biased this capacitance was negligible with respect to the wiring capacitance. However, for a low bias (0 V) the detector capacitance can be observed as an increase in the input time, Eq.(5.3). The height of the output pulses was about 20 mV; however, it depends on the output circuitry of the CCD used and is therefore only of significance within one measurement.

The lowest attainable injection time depends on the lay-out (Sec.5.2.1.). The total input capacitance has to be as low as possible (Eq.(5.8)). Because the value of the strip capacitance is fixed by the detection requirements on the detector, lower values can only be obtained by reducing the wiring capacitance. Therefore, an integrated version of the detector-CCD system was made to verify the theory [5.36]. A microphotograph of the chip is shown in Fig.5.11. Four strips, each read out by a 10-cell JCCD with an injector input, were integrated according to the principle shown in Fig.5.8a.

The four JCCDs are terminated by a parallel-in serial-out JCCD for multiplex purposes. Biasing of the injectors is accomplished by current mirrors, which are integrated between the strips and the CCDs, Fig.5.11. The strips have a pitch of 46.5 /mi, determined by the CCD layout (the gate area was 16x l6 /mi 2 ) . The first strip can be bonded, enabling "charge generation" in an electrical way.

Fig.5.11. Chip photograph of an integrated version of the detector-CCD system, with the strips on the left, the current mirrors in the middle and the CCDs on the right.

117

A conventional n-type drain makes signal output at the end of the multiplexer possible. However, a JCCD can also be read out at any gate. Therefore, some gates were not connected on chip with the clock lines, but connected separately. A gate forms, together with the n-epilayer and p-substrate, a pnp-transistor. When properly biased it can therefore be used as output, which has the additional advantage that the signal will be amplified, because of the transistor action. This vertical charge flow is also obtained by overfilling the JCCD with charge, which, for example, can be used as "anti-blooming" [5.37] or in making JCCD-logic [5.38].

First, charge packets were injected by electrical pulsing. The clock frequency was 0.8 MHz and the bias current 0.4 pA. Reading out at the end of the multiplex CCD is much more complex than reading out just at the end of the first 10 cells, because different clock pulses are needed for running the whole multiplex system. Therefore, the last gate before the multiplexer was used as output.

A capacitor of 2 pF was placed in series with the pulse generator, so that by applying wide pulses the amount of injected charge could easily be calculated. The output gate was biased through a 4 kfl resistor, across which the output pulses could be measured by an oscilloscope. The gain of the pnp output transistor could be calculated from these pulses and from the known amount of injected charge. This appeared to be about 600! This high value was confirmed by other JCCDs and test structures made on the same wafer.

Injection within one clock period could only be achieved by applying narrow pulses on the injector. Wide pulses (differentiated by the series capacitor) used at least two periods. An increase in the bias current lowered the injection time, which is in agreement with the theory of Sec.5.2.1. If the bias current is too large, charge will flow into the clock of the first gate by virtue of the "anti-blooming" mechanism. So, at the cost of dissipation and the loss of small signals the injection time can be lowered. (Bear in mind that the detector pulses are seen as a lowering of the bias current.) The "anti-blooming" effect was nicely demonstrated by a critical setting of the bias current (barely no overflow). At this setting the positive edge of the input pulse (differentiated by the series capacitor) could be seen, whereas the negative edge was invisible. (Consequently, the effect of radiation impinging on the CCD part will not be that ghost particles will be detected, but that real hits may be obscured, as experimentally could be seen from irradiating the CCD part.)

The next step was to inject charge optically. By scanning the chip with a 10 /xm wide laser beam it was seen that output could only be achieved when the laser beam hit the strip belonging to the readout CCD (or, at low bias current, when the beam hit the CCD itself).

Because short pulsing of the laser was not possible, the fast LED was used for this purpose. The CCD half of the chip was shielded for light and the LED placed above the chip. The bonding to the strip was removed, leaving a capacitance of about one pF (the strip length was 1000/xm, the pitch 46.5 fxm, and the depletion layer somewhat less than the 7 /jm epilayer

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Fig.5.12. CCD output signal (upper trace) and LED input voltage (lower trace) in the case of an integrated detector and CCD. Horizontal: 5 fis/div., vertical: 50 mV/div.

thickness). This set-up showed, Fig.5.12., that all of the charge generated by the detector was injected within one clock period, independently of the delay time after which the LED was pulsed. Taking a gain of 600 for the output transistor, the calculated injected charge is 5 fC.

In addition to the observed influence of the input capacitance and resistance (Fig.5.10.), the influence of the clock frequency was also studied. As predicted by the theory of Sec.5.2.1. it was observed that at a higher clock frequency, a larger bias current was required to regain the same settings, i.e. scaled to the new time scale (e.g. injection within one clock period, etc.). Increasing the bias current decreases the injection time (Sec.5.2.1.), thereby restoring the input time relative to the clock period. Given the charge handling capability of the CCD, at a higher clock frequency, a larger bias current is allowed (Eqs.(5.6) and (5.7)); consequently, the clock frequency and the bias current can be exchanged while preserving the same number of periods in which generated charge will be injected into the CCD.

5.3. CONCLUSIONS

The various ways of reading out a microstrip detector have been reviewed and some examples have been provided. Up to the present, all strips have been read out separately and some method of multiplexing has been used in order to reduce the number of outputs. On-chip reduction by electronic

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means is not yet feasible. The major problem prohibiting such integration stems from the compatibility demands of the detector and the electronics circuitry, which includes the material requirements, the processing, and the biasing.

All existing readout methods can be divided into two groups: passive and active readouts. The first is based on the use of some kind of resistor-capacitor network, whereas the second uses some active electronic components to modify the detector output pulses. An example of each method has been described in more detail.

Capacitive multiplexing is an example of a passive multiplex method. It is based on the use of two metalization layers, with the capacitive coupling between these layers enabling the multiplexing. The principle has been confirmed by experiments.

Readout by parallel-in serial-out CCDs belongs to the category of active multiplexers. The coupling between the detector and the CCD constitutes a critical facet, and some kind of interface is needed to match both devices to each other. The injector input is shown to enable direct coupling. The input time of this interconnection scheme has been analyzed. Complete injection of all charge generated into the CCD is only possible if the total input capacitance is related to the gate area and the charge handling capability of the CCD according to Eq.(5.8). The applicability of the method is demonstrated by experiments.

CHAPTER 6 CONCLUSIONS

This thesis deals with the silicon microstrip detector in so far as it is applied in a high-energy-physics experimental environment. Three topics concerning this device have been discussed, namely its fabrication (Chapter 3), its operation (Chapter 4) and its readout (Chapter 5). Background information was provided by Chapter 2, containing an overview of semiconductor radiation detectors. Referring to the concluding sections of each chapter (Sec.3.3., 4.3. and 5.3.), the contents of Chapter 3, 4 and 5 can be summarized as follows.

Chapter 3 deals with the fabrication of a detector. The leakage current of such a device is strongly affected by the process in which it is made. As a detector's noise contribution is mainly due to the leakage current, this current should be as low as possible, and that process resulting in the lowest leakage current should be selected for the fabrication of the detector. To achieve a satisfactory signal-to-noise ratio the detector has to be totally depleted. This ensures the collection of the entire charge generated in the detector, and thus the maximum signal level to be achieved, which still means an output signal of about only 4 fC (Sec.2.2.1.). In order to make total depletion possible the detector has to be made on high-purity silicon (Chapter 1).

For a proper interpretation of the results from leakage current measurements on actual devices, clarity about the theoretical description of this current is a prerequisite. Therefore, this theory has been reviewed in a separate section (Sec.3.1.), prior to the section presenting the experimental results of an investigation on the influence of different (standard) processes on the leakage current (Sec.3.2.).

In theory the leakage current is mainly due to generation in the depletion layer. However, diffusion current from outside this layer also contributes to the leakage current, albeit only a little bit. This is due to the fact that the generation rate, characterized by the lifetime, is" not equal in both regions. Both components can be distinguished by measuring the current's temperature dependence.

Diodes made in different (standard) processes showed the influence of the processing on the leakage current. Implantation with an anneal at 600 °C produced the best diodes. After the implantation no high temperature treatments (above 600 °C) are allowed, because they will irreparably deteriorate the diode's characteristics. However, when applied before the implantation no influence was observed. Thus, this implantation of the detector junctions should be the last step of an entire process, in which, for example, some electronics circuitry is integrated with the detector.

Chapter 4 discusses the operation of a detector. After a particle has

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generated a charge column by traversing through the detector, this charge will be collected at a few strips. Their position indicates that of the particle. The obtainable precision of such a measurement is discussed in the introductory section of Chapter 4 (Sec.4.0.1.).

During the collection process charge is induced on the actual collection strips and on their neighboring strips. Upon completion the net amount of induced charge on the former will be equal to the amount of generated charge, whereas on the latter this will be zero. The shape of the induced current pulses has been calculated (Sec.4.1.). A novel method, based upon Tellegen's theorem [4.2], is used for these calculations. The induced pulse is found by scaling all the currents flowing inside the detector due to the charge collection by some reference field and by subsequently integrating these weighted currents. The weighting field is determined separately, as it is independent of the bias conditions, etc. of the detector. Only the detector's geometry is of influence.

The calculations showed that, as expected, for the most part the currents induced in the strip which is going to collect the charge and the one next to it are equal. To end up with a total induction of zero charge, the pulse on the latter strip has a negative, compensating dip. When the junctions are at the strip side, this dip is short and sharp. The induction on the neighboring strips is the strongest when the charge is generated at the surface; it is the weakest for a charge generation throughout the whole device (as it is, for example, in the case of minimum ionizing particles).

It has been demonstrated that with strip detectors with a 20 /zm pitch a position precision of 3/mi can be obtained [4.4,4.5]. Normally, the detectors are designed with strips of 10 /im in width and with a gap of 10 /an. The effects when these sizes were varied while keeping the pitch constant were investigated (Sec.4.2.). Both theoretical and experimental results confirm each other. The main conclusion is that the position precision is only determined by the strip pitch. It is not affected by the strip width, as the electrical behavior is. A strip is more sensitive to a disturbance on its neighboring strip when the spacing between those strips is smaller. Moreover, the (dc-) resistance is lower and the interstrip capacitance larger. At higher bias voltages these effects are less pronounced.

The surface between the strips is not inverted. Therefore, the interstrip resistance is large and position measurement is possible. Otherwise there would be a channel between the strips and the precision would be lost: the generated charge would no longer be read out by one or a few strips. On the other hand, an accumulated surface enhances the recombination of generated electron holes. In addition, the (ac-)impedance is influenced by the presence of (free) charge carriers at the surface.

Chapter 5 presents the readout possibilities of a detector. The pitch of the strips in a microstrip detector is typically 20 pm, while the device itself is of wafer size. This means that a few thousand strips have to be read out, and consequently, some multiplexing has to be done. The existing readout

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schemes, classified into passive and active schemes, have been summarized (Sec.5.1.). The former are based upon resistor-capacitor networks, while the latter use some kind of active components. One example of each has been outlined a little bit further. The compatibility demands for the implementation of the active readouts on the same wafer as that on which the detector is made have been discussed (Sec.5.1.2.).

Capacitive multiplexing, making use of a second metalization layer, is an example of a passive readout (Sec.5.1.1.). It is based on the principle of subdividing the strips into groups, the combined strip and group number then indicating the position of an entering particle. The first strips of each group are connected to each other in the first metalization layer, as are the second, the third, and so on. Each group is covered by a plate made in the second metalization layer. When a particle hits a strip not only that strip, but also the plate above it gives an output pulse, because of the capacitive coupling between them.

The use of JCCDs is an example of an active readout (Sec.5.2.). One of the problems encountered in realizing such a readout is brought about by the coupling between the detector and the CCD; the detector has a large capacitance, while the usual CCD input behaves as a large resistance, and consequently the charge transport from the detector into the CCD will take several clock cycles. The injector input of the JCCD overcomes this problem: all of the collected charge on a strip can be injected into the CCD within a few clock cycles, provided that the total amount of charge needed to charge the input capacitance to a few kT/q is smaller than the maximum amount of charge that the CCD can transport in one clock potential well (into which, at all events, the total amount of collected charge should naturally fit).

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CHAPTER 2 OVERVIEW

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[3.29] H. Ryssel and I. Ruge, Ionenimplantation, B.G. Teubner, Stuttgart, 1978

[3.30] J. Kemmer, P. Burger, R. Henck and E. Heijne, Performance and applications of passivated ion-implanted silicon detectors, IEEE Trans. Nucl. Sci. NS-29 (1982) 733-737

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1041-1051 [3.34] P. Eichinger, Characterization and analysis of detector materials

and processes, Nucl. Instr. & Meth. A253 (1987) 313-318 [3.35] P. Zandveld, Some properties of ion-implanted p-n junctions in

silicon, Solid-St. Electron. 19 (1976) 659-667 [3.36] S. Oosterhoff and J. Middelhoek, The annealing of I MeV

implantations of boron in silicon, Solid-St. Electron. 28 (1985) 427-433

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[3.38] J. Buechler, E. Kasper, P. Russer and K.M. Strohm, Silicon high-resistivity-substrate millimeter-wave technology, IEEE Trans. Electr. Dev. ED-33 (1986) 2047-2052

[3.39] H.Y. Tsoi, J.P. Ellul, M.I. King, J.T. White and W.C. Bradley, A deep-depletion CCD imager for soft X-ray, visible, and near-infrared sensing, IEEE Trans. Electr. Dev. ED-32 (1985) 1525-1530

[3.40] M. Vos, D.O. Boerma, P.J.M. Smulders and S. Oosterhoff, Defect and dopant depth profiles in boron-implanted silicon studied with channeling and nuclear reaction analysis, Nucl. Instr. & Meth. B17 (1986) 234-241

CHAPTER 4 OPERATION

[4.1] W.R.Th, ten Kate, Applying Tellegen's theorem to calculate the pulse responses on a microstrip detector's strips, IEEE Trans. Nucl. Sci. NS-34 (1987)

[4.2] P. Penfield, R. Spence and S. Duinker, Tellegen's theorem and electrical networks, The M.I.T. Press, Cambridge (Mass.), 1970

[4.3] W.R.Th, ten Kate, The influence of the strip width on the perfor­mance of a strip detector, Nucl. Instr. & Meth. A253 (1987) 333-349

[4.4] R. Klanner, Silicon detectors, Nucl. Instr. & Meth. A235 (1985) 209-215

[4.5] E. Belau, R. Klanner, G. Lutz, E. Neugebauer, H.J. Seebrunner, A. Wylie, T. Bohringer, L. Hubbeling, P. Weilhammer, J. Kemmer, U. Kötz and M. Riebesell, Charge collection in silicon strip detectors, Nucl. Instr. & Meth. 214 (1983) 253-260

[4.6] P.F. Manfredi and F. Ragusa, Microvertex detectors, present trends and future perspectives, Nucl. Instr. & Meth. A252 (1986) 208-226

[4.7] C.J.S. Damerell, Developments in solid state vertex detectors, lectures presented at the SLAC summer institute, RAL-84-123 , 1984

[4.8] R. Hofmann, R. Klanner, G. Lutz, G. Lütjens, A. Wylie and J. Kemmer, High resolution silicon detectors - status and future developments of the MPl/TU-Munich group, Nucl. Instr. & Meth. 225(1984)601-605

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[4.9] W. Seibt, K.E. Sundström and P.A. Tove, Charge collection in silicon detectors for strongly ionizing particles, Nucl. Instr. & Meth. 113 (1973)317-324

[4.10] E. Gatti, A. Longoni, P. Rehak and M. Sampietro, Dynamics of electrons in drift detectors, Nucl. Instr. & Meth. A253 (1987) 393-399

[4.11] H.W. Kraner, R. Beuttenmuller, T. Ludlam, A.L. Hanson, K.W. Jones, V. Radeka and E.H.M. Heijne, Charge collection in silicon strip detectors, IEEE Trans. Nucl. Sci. NS-30 (1983) 405-414

[4.12] I. Duerdoth, Track fitting and resolution with digital detectors, Nucl. Instr. & Meth. 203 (1982) 291-297

[4.13] S. Ramo, Currents induced by electron motion, Proc. IRE 27 (1939) 584-585

[4.14] G. Cavalleri, E. Gatti, G. Fabri and V. Svelto, Extension of Ramo's theorem as applied to induced charge in semiconductor detectors, Nucl. Instr. & Meth. 92 (1971) 137-140

[4.15] J.B. Gunn, A general expression for electrostatic induction and its application to semiconductor devices, Solid-St. Electron. 7 (1964) 739-742

[4.16] E. Gatti, G. Padovini and V. Radeka, Signal evaluation in multi-electrode radiation detectors by means of a time dependent weighting vector, Nucl. Instr. & Meth. 193 (1982) 651-653

[4.17] W.R. Smythe, Static and dynamic electricity, 3 r d ed., McGraw-Hill, New York, 1968

[4.18] K.J. Binns and P.J. Lawrenson, Analysis and computation of electric and magnetic field problems, 2 n d ed., Pergamon Press, Oxford, 1973

[4.19] H.F. Wolf, Silicon semiconductor data, Pergamon Press, Oxford, 1969

[4.20] R.N. Williams and E.M. Lawson, Formation of current pulses in semiconductor nuclear radiation detectors, Nucl. Instr. & Meth. 113 (1973)597-598

[4.21] A.S. Grove, Physics and technology of semiconductor devices, John Wiley & Sons, New York, 1967

[4.22] M.S. Adler, V.A.K. Temple, A.P. Ferro and R.C. Rustay, Theory and breakdown voltage for planar devices with a single field limiting ring, IEEE Trans. Electr. Dev. ED-24 (1977) 107-113

[4.23] S.M. Sze, Physics of semiconductor devices, John Wiley & Sons, New York, 1969

[4.24] J.T. Walton and F.S. Goulding, Silicon radiation detectors with oxide charge state compensation, IEEE Trans. Nucl. Sci. NS-34 (1987) 396-400

[4.25] W. Fichtner, D.J. Rose and R.E. Bank, Semiconductor device simulation, IEEE Trans. Electr. Dev. ED-30 (1983) 1018-1030

[4.26] S. Lerose, D. Pons and C. Fonné, Microslrip detectors with inverted structure for hardening against radiation damage, IEEE Trans. Nucl. Sci. NS-33 (1986) 347-350

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[4.27] V. Boisson, M. Le Helley and J.P. Chante, Analytical expression for the potential of guard rings of diodes operating in the punchthrough mode, IEEE Trans. Electr. Dev. ED-32 (1985) 838-840

[4.28] E.H. Nicollian and A. Goetzberger, The Si-Si02 interface -electrical properties as determined by the metal-insulator-silicon conductance technique, Bell Syst. Tech. J. 46 (1967) 1055-1133

[4.29] S.P. Kwok, An X-band SOS resistive gate-insulator-semiconductor (RIS) switch, IEEE Trans. Electr. Dev. ED-27 (1980) 442-448

[4.30] U. Kötz, K.U. Pösnecker, E. Gatti, E. Belau, D. Buchholz, R. Hofmann, R. Klanner, G. Lutz, E. Neugebauer, H.J. Seebrunner, A. Wylie and J. Kemmer, Silicon strip detectors with capacitive charge division, Nucl. Instr. & Meth. A235 (1985) 481-487

CHAPTER 5 READOUT

[5.1] P. Jarron and M. Goyot, A fast current sensitive preamplifier (MSD2) for the silicon microstrip detector, Nucl. Instr. & Meth. 226 (1984) 156-162

[5.2] W. Buttler, B.J. Hosticka, G. Lutz and G. Zimmer, Low power-low noise monolithic detector readout electronics, Nucl. Instr. & Meth. A253 (1987)439-443

[5.3] W.R.Th, ten Kate and S.A. Audet, Processing high-purity silicon used for sensor applications, Proc. 4 t h International Conference on Solid-State Sensors and Actuators, Transducers '87, Tokyo, June 1987, Digest of Technical Papers, 103-106

[5.4] B.J. Hosticka and G. Zimmer, Integration of detector arrays and readout electronics on a single chip, IEEE Trans. Nucl. Sci. NS-32 (1985)402-408

[5.5] J.T. Walker, S. Parker, B. Hyams and S.L. Shapiro, Development of high density readout for silicon strip detectors, Nucl. Instr. & Meth. 226 (1984)200-203

[5.6] R.W. Johnson, J.L. Davidson, R.C. Jaeger and D.V. Kerns, Silicon hybrid wafer-scale package technology, IEEE J. Solid-St. Circuits SC-21(1986) 845-851

[5.7] R. Klanner, Silicon detectors, Nucl. Instr. & Meth. A235 (1985) 209-215

[5.8] C.J.S. Damerell, Developments in solid state vertex detectors, lectures presented at the SLAC summer institute, RAL-84-123 , 1984

[5.9] P.F. Manfredi and F. Ragusa, Microvertex detectors, present trends and future perspectives, Nucl. Instr. & Meth. A252 (1986) 208-226

[5.10] W.R.Th, ten Kate, Detectors for nuclear radiation, Sensors & Actuators 10 (1986) 83-101

[5.11] E. Laegsgaard, Position-sensitive semiconductor detectors, Nucl. Instr. & Meth. 162 (1979) 93-111

[5.12] J.E. Lamport, G.M. Mason, M.A. Perkins and A.J. Tuzzolino, A large area circular position sensitive Si detector, Nucl. Instr. & Meth. 134

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(1976)71-76 [5.13] V. Radeka, Signal, noise and resolution in position-sensitive

detectors, IEEE Trans. Nucl. Sci. NS-21 (1974) 51-64 [5.14] J.L. Alberi and V. Radeka, Position sensing by charge division, IEEE

Trans. Nucl. Sci. NS-23 (1976) 251-258 [5.15] J.B.A. England, B.D. Hyams, L. Hubbeling, J.C. Vermeulen and P.

Weilhammer, Capacitative charge division read-out with a silicon strip detector, Nucl. Instr. & Meth. 185 (1981) 43-47

[5.16] U. Kötz, K.U. Pösnecker, E. Gatti, E. Belau, D. Buchholz, R. Hofmann, R. Klanner, G. Lutz, E. Neugebauer, H.J. Seebrunner, A. Wylie and J. Kemmer, Silicon strip detectors with capacitive charge division, Nucl. Instr. & Meth. A235 (1985) 481-487

[5.17] E. Gatti, A. Longoni, R.A. Boie and V. Radeka, Analysis of the position resolution in centroid measurements in MWPC, Nucl. Instr. & Meth. 188 (1981)327-346

[5.18] N.J. Hansen, D.J. Henderson and R.G. Scott, A position sensitive surface barrier array type detector, Nucl. Instr. & Meth. 105 (1972) 293-300

[5.19] W.R.Th, ten Kate and H.M. Heijne, Capacitive multiplexing on a silicon microstrip detector, Nucl. Instr. & Meth. A234 (1985) 398-400

[5.20] Ch. Adolphsen, A. Litke, A. Schwarz, M. Turala, V. Liith, A. Breakstone and S. Parker, Test beam results for silicon microstrip detectors with VLSI readout, Nucl. Instr. & Meth. A253 (1987) 444-449

[5.21] S.A. Audet and W.R.Th, ten Kate, Two-dimensional silicon sensor for imaging radiation, Proc. 4 t h International Conference on Solid-State Sensors and Actuators, Transducers '87, Tokyo, June 1987, Digest of Technical Papers, 267-270

[5.22] R. Alberganti, E. Chesi, Ch. Gerke, F. Piuz, L. Ramello, T.D. Williams and R. Roosen, A system of 4400 silicon microstrips readout with analog multiplexed electronics used in the WA75 experiment, Nucl. Instr. & Meth. A248 (1986) 337-353

[5.23] P .E : Karchin, D.L. Hale, R.J. Morrison, M.S. Witherell, M.D. Sokoloff, A. Kiang, B.R. Kumar, J.F. Martin and M. Sarabura, Test beam studies of a silicon microstrip vertex detector, IEEE Trans. Nucl. Sci. NS-32 (1985) 612-615

[5.24] H. Anders, J. Feyt, E. Heijne, M. Iten, P. Jarron, P. Montanari, U. Spadinger, R. Zurbuchen, D. Daub and H. Effing, Feasibility study for a charge coupled device (CCD) as a multiplexing signal processor for microstrip and other particle detectors, CERN/EF 85-10, 1985

[5.25] R. Hofmann, G. Lutz, B.J. Hosticka, M. Wrede, G. Zimmer and J. Kemmer, Development of readout electronics for monolithic integration with diode strip detectors, Nucl. Instr. & Meth. 226 (1984)196-199

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[5.26] H.Y. Tsoi, J.P. Ellul, M.I. King, J.T. White and W.C. Bradley, A deep-depletion CCD imager for soft X-ray, visible, and near-infrared sensing, IEEE Trans. Electr. Dev. ED-32 (1985) 1525-1530

[5.27] G. Zimmer and H. Vogt, CMOS on buried nitride-a VLSI SOI technology, IEEE Trans. Electr. Dev. ED-30 (1983) 1515-1520

[5.28] W.R.Th, ten Kate and C.L.M. van der Klauw, A new readout structure for radiation silicon strip detectors, Proc. 29 t h IEDM, Washington DC, Dec. 1983, IEDM Technical Digest, 647-650

[5.29] W.R.Th, ten Kate and C.L.M. van der Klauw, Strip detector - CCD coupling by means of a bipolar transistor, Nucl. Instr. & Meth. 226 (1984)193-195

[5.30] W.S. Chan, Detector-charge-coupled device (CCD) interface methods, Proc. Soc. Photo-Opt. Instr. Eng. 244 (1980) 81-96

[5.31] G.S. Hobson, Charge-transfer devices, Edward Arnold, London, 1978

[5.32] L.J.M. Esser, Charge coupled devices: physics, technology and applications, Tijdschr. NERG 47 (1982) 51-64

[5.33] C.H. Séquin and M.F. Tompsett, Charge transfer devices, Academic Press., New York, 1975

[5.34] M. Kleefstra, First experimental bipolar Charge-Coupled Device, Microelectr. 7 (1975) 68-69

[5.35] M. Kleefstra and E.A. Wolsheimer, Junction Charge-Coupled Devices, Proc. 25 t h IEDM, Washington DC, Dec. 1979, IEDM Technical Digest, 615-618

[5.36] W.R.Th, ten Kate and C.L.M. van der Klauw, Experimental results on an integrated strip detector with CCD readout, Nucl. Instr. & Meth. 228 (1984) 105-109

[5.37] CD. Hartgring and M. Kleefstra, Quantum efficiency and blooming suppression in Junction Charge-Coupled Devices, IEEE J. Solid-St. Circuits SC-13 (1978) 728-730

[5.38] E.P. May, C.L.M. van der Klauw, M. Kleeftsra and E.A. Wolsheimer, Junction Charge-Coupled Logic (JCCL), IEEE J. Solid-St. Circuits SC-18 (1983) 767-772

SAMENVATTING DE SILICIUM MICROSTRIPDETECTOR Dit proefschrift behandelt de silicium microstripdetector zoals die

gebruikt wordt voor experimenten in de hoge-energiefysica. Drie aspecten van de detector worden in het proefschrift besproken, namelijk zijn fabricage (Hoofdstuk 3), zijn werking (Hoofdstuk 4) en zijn uitlezing (Hoofdstuk 5). In het inleidende hoofdstuk, Hoofdstuk 2, wordt meer algemeen ingegaan op de halfgeleiderstralingsdetectoren.

In Hoofdstuk 3 wordt de fabricage van de microstripdetector behandeld. De lekstroom hangt in hoge mate af van het proces waarin de diode gemaakt is. Daar de ruisbijdrage van een detector voornamelijk een gevolg is van de lekstroom, moet deze stroom zo laag mogelijk zijn. Het proces dat in de laagste lekstroom resulteert, moet dus voor de fabricage van de detector uitgekozen worden. Voor een redelijke signaal-ruis verhouding is het echter ook nodig de detector geheel te depleteren. Hierdoor wordt het mogelijk alle lading die in de detector gegenereerd is, te collecteren, met als resultaat dat het maximaal haalbare signaal niveau bereikt kan worden. Dat niveau betekent trouwens nog steeds slechts een signaal van ongeveer 4 fC. Om een detector geheel te depleteren, is het noodzakelijk dat hij op hoogohmig silicium gemaakt wordt.

Voor een juiste interpretatie van lekstroommetingen aan gemaakte detectoren mag er geen onzekerheid over de theoretische beschrijving van de lekstroom bestaan. Het hoofdstuk geeft daarom eerst een overzicht van die theorie, alvorens in te gaan op de experimentele resultaten van een onderzoek naar de invloed van de verschillende (standaard)processen op de lekstroom.

Theoretisch is de lekstroom voornamelijk een gevolg van generatie in de depletielaag. De diffusiestroom buiten deze laag draagt echter ook bij, al is het maar voor een klein deel. Dit komt omdat de generatiesnelheid, gekarakteriseerd door de levensduur, in beide delen niet even groot is. Door middel van hun temperatuurafhankelijkheid kunnen de beide componenten onderscheiden worden.

Diodes gemaakt in verschillende (standaard)processen toonden de proces­invloed op de lekstroom. Implantatie met een 600 °C-anneal produceerde de beste diodes. Na de implantatie zijn geen hoge temperatuurbehandelingen (boven 600 C) meer toegestaan, omdat die de diodekarakteristieken onherstelbaar slechter maken. Vinden deze behandelingen echter voor de implantatie plaats, dan is geen invloed merkbaar. In een compleet proces, waarin b.v. enige electronica met de detector wordt geïntegreerd, moet de implantatie van de detectorjuncties dus de laatste stap zijn.

In Hoofdstuk 4 wordt de werking van de microstripdetector bekeken. Na­dat een passerend deeltje een ladingskolom in de detector heeft gegenereerd,

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zal deze lading door enkele strips gecollecteerd worden. Hun positie geeft die van het deeltje aan. De inleidende paragraaf van Hoofdstuk 4 bespreekt de bereikbare precisie van zo'n meting.

Gedurende de collectie zal er zowel lading geïnduceerd worden op de uiteindelijke collectiestrips als op de daaraan aangrenzende strips. Ui t ­eindelijk zal de netto geïnduceerde hoeveelheid lading op de eerste gelijk zijn aan de gegenereerde hoeveelheid lading, terwijl die op de laatste nul zal zijn. De vorm van de geïnduceerde stroompulsen is berekend. Voor deze berekening is gebruik gemaakt van een nieuwe methode, gebaseerd op de stelling van Tellegen [4.2]. De geïnduceerde puls wordt gevonden door alle stromen die binnenin de detector vloeien ten gevolge van de ladingscollectie, te schalen met een referentieveld en door vervolgens deze gewogen stromen te integreren. Het weegveld, dat onafhankelijk is van de biascondities enz. van de detector, wordt apart bepaald (alleen de geometrie van de detector is van invloed).

De berekeningen toonden dat, zoals verwacht, de stromen geïnduceerd in de strip die uiteindelijk de lading zal collecteren en geïnduceerd in diens buurstrip voor het grootste deel aan elkaar gelijk zijn. Aangezien voor de laatste het totaal van de geïnduceerde lading tenslotte op nul moet uitkomen, heeft daar de puls een negatieve, compenserende dip. Als de juncties aan de kant van de strips gevormd zijn, zal deze dip kort en scherp zijn. De inductie op de buurstrips is het sterkst als de lading aan het oppervlak is gegenereerd, terwijl zij het zwakst is bij een' ladingsgeneratie over het hele device (zoals b.v. in het geval van minimum ionizing particles).

Anderen hebben aangetoond dat met stripdetectors met een 20 /im pitch een precisie in de plaatsbepaling van 3 pm gehaald kan worden [4.4,4.5]. Gewoonlijk worden detectors ontworpen met strip- en gapbreedtes van lO^tm. De gevolgen van een variatie in deze afmetingen bij een constante pitch zijn onderzocht, zowel theoretisch als experimenteel. De resultaten waren in overeenstemming met elkaar. De voornaamste conclusie is dat de positieprecisie alleen van de pitch afhangt. Zij wordt niet beïnvloed door de stripbreedte, zulks in tegenstelling tot het elektrische gedrag. Een strip is gevoeliger voor een verstoring op zijn buurstrip als de ruimte tussen de strips kleiner wordt. Bovendien is dan de (dc-)weerstand kleiner en de interstrip-capaciteit groter. Bij een hogere biasspanning zijn deze effecten minder geprononceerd.

Het oppervlak tussen de strips is niet geïnverteerd. Hierdoor blijft de interstripweerstand hoog en is positiemeting mogelijk. Anders zou er een kanaal tussen de strips zijn en de precisie verloren gaan: de gegenereerde lading zou niet langer door één of enkele strips uitgelezen worden. Een geaccumuleerd oppervlak anderzijds, vergroot de recombinatie van gegenereerde elektronen en gaten. Daarnaast wordt de (ac-)impedantie beïnvloed door de aanwezigheid van (vrije) ladingdragers aan het oppervlak.

In Hoofdstuk 5 wordt de uitlezing van de microstripdetector besproken. De strippitch is typisch 20 /xm, terwijl het device zelf een hele wafer beslaat. Dit betekent dat een paar duizend strips uitgelezen moeten worden en dus dat

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er multiplexing nodig is. Een resumé van de bestaande uitleesmethoden, onder te verdelen in passieve en actieve methoden, wordt in het hoofdstuk gegeven. De eerste zijn gebaseerd op weerstand-capaciteit netwerken, terwijl de laatste gebruikmaken van actieve componenten. Van beide groepen is er een voorbeeld in iets meer detail uitgewerkt. De compatibiliteitseisen om de actieve uitlezing op dezelfde wafer te implementeren als waarop de detector is gemaakt, zijn besproken.

De capacitieve-multiplexing methode, gebaseerd op het gebruik van een tweede metallisatie laag, is een vertegenwoordiger van de passieve uitlees­methoden. Het principe berust op het onderverdelen van de strips in groepen, waarbij dan de combinatie van strip- en groepnummer de positie van een invallend deeltje aangeeft. De eerste strips van elke groep worden met elkaar verbonden in de eerste metallisatie laag, net als de tweede, de derde, enz. Elke groep wordt bedekt door een plaat gemaakt in de tweede metallisatie laag. Als een deeltje nu een strip raakt, zal niet alleen die strip, maar ook de plaat erboven vanwege de capacitieve koppeling een puls geven.

Het gebruik van JCCD's is een voorbeeld van een actieve uitleesmethode. Eén van de problemen om deze uitlezing te realiseren, vormt de koppeling tussen detector en CCD (de detector heeft een grote capaciteit, terwijl de gebruikelijke CCD-ingang zich als een hoge weerstand gedraagt; het ladingstransport van de detector in de CCD zal dus enkele klokcycli duren). De injectoringang van de JCCD verhelpt dit probleem: alle door een strip gecollecteerde lading kan in de CCD geïnjecteerd worden binnen een paar klokcycli. Voorwaarde hierbij is dat de totale hoeveelheid lading die nodig is om de ingangscapaciteit tot enkele kT/q op te laden kleiner is dan de maximale hoeveelheid lading die de CCD kan transporteren in één poten-tiaalput (waarin uiteraard in ieder geval de totale hoeveelheid gecollecteerde lading moet passen).

Het proefschrift besluit met Hoofdstuk 6, waarin de conclusies van de voorgaande hoofdstukken nog eens worden samengevat.

ACKNOWLEDGEMENT

This thesis could not be written without the stimulating interest and enthusiasm I have experienced from all the people around me. This is equal in value to the assistance I received in the actual realization of this thesis. I would like to express my thanks by acknowledging the following few persons, in particular.

For my involvement and education in semiconductor physics I am indebted to Prof. Dr. S. Middelhoek, who finally led me to the writing of this thesis. I am very grateful to him for the fatherly way he offered help and stimulation, as well as his willingness to listen to all of my problems and discuss them any time.

Thanks to Ms. S. Massotty I have been able to verbalize my thoughts. She patiently read my manuscript and changed it to a correct English format. My warmest gratitude is extended to her.

I am greatly obliged to Dr. H.M. Heijne, who introduced the microstrip detector to our laboratory, for his help, for the many fruitful discussions we had, and last but not least, for his hospitality during my visits to CERN, which he also arranged for me.

Many devices have been designed to perform experiments. They were fabricated at the IC-Workshop of the Delft University of Technology, except for the capacitive multiplex and integrated detector-CCD devices. These were processed at Philips Industries in Nijmegen, for which I would like to express my appreciation. Often, different (non-standard) processes had to be applied to the wafers within one batch. I greatly admire the unfailing patience displayed to obtain good quality devices and I am deeply indebted to E.J.G. Goudena, J. Groeneweg, F.J. de Jong, W. de Koning, E.J. Linthorst, P.K. Nauta, E. Smit, J.M.G. Teven, W. Verveer and L. Wubben.

I received valuable contributions from the work carried out by students working on this project, not the least of which were their critical remarks. In the first place, I would like to thank P.J. van Wijnen, not only for being the agreeable person he is, but also for his never-ending perseverance, which led to an increasing knowledge of the detector. Likewise, I wish to express my gratitude to D.W. de Bruin and J. van der Graaf.

Without the support from scientific and technical staff members this thesis could never be realized. I am indebted to Prof. Dr. M. Kleefstra for the interest and help he showed in my work, especially concerning the JCCD part. It is a pleasure to be able to express my thanks to Prof. Dr. A.T. de Hoop, who pointed out the use of Tellegen's theorem for the calculation of

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the pulse responses. I will remember the pleasant evenings we had playing music. The discussions I had with the staff members of the laboratory has certainly improved the thesis. Equally important is the assistance I received during my struggles with apparatus. I would like to thank M.J. Geerts, Ms. P.M. Gerlach, G. de Graaf, Ms. C. van der Hout, Dr. J.H. Huijsing, A. van der Male, Dr. P.P.L. Regtien, F. Schneider, J.C. Staalenburg, J.B. van Staden, R.A. Stolk, Dr. M. Stuivinga, P.J. Trimp, M.J. Vellekoop and Dr. A. Venema.

I was fortunate to receive support from many sides. I wish to acknowledge Dr. R.W. Hollander and E.M. Schooneveld of the I.R.I, for their enthusiasm and help in testing the detectors with radiation, as well as for the fruitful discussions we had about the measurement results; R. Bruinink and J.M. Koopmans for always putting their measurement apparatus at my disposal; and A.A. Baas, P.J.J. van Daalen, S. de Graaf, M. Quispel and B. Zorn for their assistance with the departmental computer facilities.

All the figures in this thesis were excellently drawn by the department's technical-drawing staff. It is a pleasure to be able to express my appreciation to G. van Berkel, J.W. Muilman and W.J.P. van Nimwegen, not only for making the drawings in this thesis, but also for all the other drawings needed for publications. This applies equally well to the department's photographers, J.C. van der Krogt and J.C. Schipper, who made photoprints, copyproofs, overheads and slides for me, with great craftsmanship.

It is impossible to thank everybody without leaving somebody out. It is likewise impossible to list each person's (moral) support in the first place. At the end of this acknowledgement, I would like to mention without name my fellow students, friends, relatives and family, who surely know that they are not the last to be remembered by me.

BIOGRAPHY

Warner ten Kate was born in Leiden, The Netherlands, on June 6, 1959. He studied electrical engineering at the Delft University of Technology, and graduated in 1982 (cum laude). For this study he was awarded the 1983-prize of the Delft University Fund. As a graduate student he carried out research on amorphous-silicon solar cells and on silicon radiation detectors. He is the author of several technical papers on the latter subject, which constitutes the subject of this thesis as well. Some of these have been presented at international conferences in Europe, the U.S.A. and Japan. Starting 1988 he will join the Philips Research Laboratories in Eindhoven, The Netherlands.

Being a fervent amateur player of the French Horn, Warner has played in many orchestras, including the Dutch University Orchestra and the National Youth Orchestra. He has also performed several times as a soloist. In 1985 he began studying at the Royal Conservatory of Music, The Hague, The Netherlands, where he will take his final exams in June 1989.

STELLINGEN

behorende bij het proefschrift The Silicon Microstrip Detector

Warner ten Kate

I

Het is mogelijk om met behoud van detectorkwaliteit electronica en detector op eenzelfde wafer te integreren.

Dit proefschrift, Hoofdstuk 3.

n Met behulp van de stelling van Tellegen kan een eenvoudige uitdrukking afgeleid worden voor de berekening van de pulsvormen op de strips van een microstripdetector ten gevolge van ladingscollectie in die detector.

Dit proefschrift, Hoofdstuk 4.

m De JCCD- injector inlaat kan succesvol gebruikt worden om een stripdetector rechtstreeks aan een CCD te koppelen. Dit proefschrift. Hoofdstuk 5.

IV

Een tweede metallisatielaag zoals toegepast in de in het proefschrift beschreven capacitieve-multiplexing methode, kan met succes gebruikt worden voor het vervaardigen van een tweedimensionale-positiedetector.

V

De "packaging" vormt een wezenlijk aspect van een sensor. Zij verdient dan ook meer aandacht dan tot op heden door het sensoronderzoek aan haar gegeven is.

VI

De terughoudendheid waarmee waarnemingen van kunstmatige sensoren worden benaderd, staat in schril contrast met het gemak waarmee de juist zeer subjectieve zintuigelijke waarnemingen worden geaccepteerd.

VII

Als de theoretische natuurkunde ernaar streeft alle natuurkrachten tot één fundamentele te herleiden ("unificatietheorie"), dan zullen op z'n minst ook verklaringen gegeven moeten worden voor metafysische verschijnselen.

VIII

Alle waarnemingen zijn relatief, zelfs het absolute gehoor.

IX

Muziek voor 1820 gecomponeerd komt beter tot haar recht indien zij op (kopieën van) authentieke instrumenten ten gehore wordt gebracht.

X

De taal is zeer redundant; dit blijkt vooral tijdens vergaderingen.

XI

In de politieke discussie gaat het veelal niet over welke weg, maar of deze rechts dan wel links bewandeld moet worden.

XII

Gezien de huidige maatschappijstructuur zouden cursussen als typen en autorijles standaard in het onderwijspakket opgenomen moeten worden.

XIII

Een wetenschappelijke stelling die niet op eigen onderzoek berust, kan nooit wetenschappelijk zijn. Promotiereglement Technische Universiteit Delft, art.5.2.: Aan het proefschrift worden ten minste zes niet op het onderwerp betrekking hebbende, wetenschappelijk verantwoorde en verdedigbare stellingen toegevoegd.

XIV

Gezien de verscheidenheid in definities en meetmethoden is er aan het resultaat van een ladingdragerslevensduurmeting veelal geen r vast te knopen.